FAIRCHILD LED Application Design Guide Using Half-Bridge LLC Resonant Converter for 100W Street Lighting handbook
Contents
1. Shutdown Figure 16 Functional Block Diagram of FSFR Series 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 www fairchildsemi com AN 9729 C 2N2907 Recor Dacor S z Civec R102 10K W 10k T pe Gace V ali R103 Miva 6 8V a CBooT Rmax Rain Rss i I VCTR Css i il RN D 3 RseENSE APPLICATION NOTE D1 2N2222 C201 R202 R201 Figure 17 Reference Circuit for Design Example of LLC Resonant Half Bridge Converter Design Procedure In this section a design procedure is presented using the schematic in Figure 17 as a reference An integrated transformer with center tap secondary side is used and input is supplied from Power Factor Correction PFC pre regulator A DC DC converter with 100W 100V output has been selected as a design example The design specifications are as follows Nominal input voltage 400Vpc output of PFC stage Output 100V 1A 100W Hold up time requirement 30ms 50Hz line freq DC link capacitor of PFC output 240uF STEP 1 Define System Specifications Estimated Efficiency E The power conversion efficiency must be estimated to calculate the maximum input power with a given maximum output power If no reference data is available use Eg 0 88 0 92 for low voltage output applications and Ey 0 92 0 96 for high voltage output applications With the estimated e2. Q 1 00 Q 0 75 e Q 0 50 gt qi Q 0 25 wW S v SS Frequency kHz Figure 9 Typical Gain Curves of LLC Resonant Converter m 3 Using an Integrated Transformer Consideration of Operation Mode and Attainable Maximum Gain Operation Mode The LLC resonant converter can operate at frequency below or above the resonance frequency f as illustrated in Figure 10 Figure 11 shows the waveforms of the currents in the transformer primary side and secondary side for each operation mode Operation below the resonant frequency case I allows the soft commutation of the rectifier diodes in the secondary side while the circulating current is relatively large The circulating current increases more as the operation frequency moves downward from the resonant frequency Meanwhile operation above the resonant frequency case II allows the circulating current to be minimized but the rectifier diodes are not softly commutated Below resonance operation is preferred for high output voltage applications such as street LED lighting systems where the reverse recovery loss in the rectifier diode is severe Below resonance operation has a narrow frequency range with respect to the load variation since the frequency is limited below the resonance frequency even at no load condition On the other hand above resonance operation has less conduct
3. APPLICATION NOTE Table 1 Pin Description This pin is the drain of the high side MOSFET typically connected to the input DC link voltage 1 This pin is for discharging the external soft start capacitor when any protections are triggered When the voltage of this pin drops to 0 2V all protections are reset and the controller starts to operate again owe This pin is to program the switching frequency Typically opto coupler and 3 RT l ge resistor are connected to this pin to regulate the output voltage ale ps sc ce re This pin is to sense the current flowing through the low side MOSFET Typically negative voltage is applied on this pin This pin is the control ground This pin is the power ground This pin is connected to the source of the low side MOSFET This pin is the supply voltage of the control IC No connection c HV This pin is the supply voltage of the CC high side drive circuit This pin is the drain of the low side 10 VcTR MOSFET Typically transformer is connected to this pin 4 5 Internal I Bias S LUV LUV HUV HUV High Side Gate Driver Level Shifter Divider Low Side Gate Driver Balancing Delay
4. ba r R sense Ns Ret EREE Figure 27 Full Wave Sensing Figure 28 Example of CC and CV Feedback Circuit 2011 Fairchild Semiconductor Corporation www fairchildsemi com Rev 1 0 0 3 22 11 13 AN 9729 Figure 29 shows another example of a CC and Over Voltage Regulation OVR Mode feedback circuit for multi output LED power supply The FAN7346 is a LED current balance controller that controls four LED arrays to maintain equal LED current To prevent LED driving voltage being over the withstanding voltage of component the FAN7346 controls LED driving voltage The OVR control circuit activates when the ENA pin is in HIGH state If OVR pin voltage is lower than 1 5V the Feedback Control FB pin voltage follows headroom control to maintain minimum voltage of drain voltages as 1V If OVR pin voltage is higher than 1 5V the FAN7346 controls FB FB is pulled LOW through FB regulation so the OVR pin voltage is not over 1 5V LED current is controlled by FBx pin voltage The external current balance switch is operating in linear region to control LED current Sensed voltage at the FBx pin is compared with internal reference voltage and controller signals the gate or base for external current balance switch Internal reference voltage is made from ADIM voltage The LED current is determined as V ADIM I ___ADIM LED Toy Reauge 35 ADIM voltage is clamped internally from 0 5V to 4V The protections such as ope
5. lao 1s obtained as ac Ti sin at 1 and Vp 1s given as Va tV if sin t gt 0 0 Va V if sin at lt 0 where V is the output voltage The fundamental component of Vz is given as Vi Z sin t 3 Since harmonic components of Vp are not involved in the power transfer AC equivalent load resistance can be calculated by dividing Ve by Inc as Vn es Ri n T 4 Considering the transformer turns ratio n N N the equivalent load resistance shown in the primary side is obtained as 2 a i R 5 IC By using the equivalent load resistance the AC equivalent circuit is obtained as illustrated in Figure 6 where V and Vro are the fundamental components of the driving voltage V and reflected output voltage Vro NV py respectively Figure 5 Derivation of Equivalent Load Resistance Rac 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE Figure 6 AC Equivalent Circuit for LLC Resonant Converter With the equivalent load resistance obtained in Equation 5 the characteristics of the LLC resonant converter can be derived Using the AC equivalent circuit of Figure 6 the voltage gain M is obtained as 6 m 1 O CE 0 where Q Io TOS jon As can be seen in Equation 6 there are two resonant frequencies One is determined by L and C while the other is determined by L and C Equation 6
6. Design Example The voltage stress and current stress of _ the rectifier diode are Vp 2V Vr 2 100 0 9 201 8V D 7 1 0 7854 The 600V 8A Ultra fast recovery diode is selected for the rectifier considering the voltage overshoot caused by the _ stray inductance The RMS current of the output capacitor is T Eee oy RMS BoP ai I 0 48A4 C AE o g o When two electrolytic capacitors with ESR of 100mQ are used in parallel the output voltage ripple is given as AV I Re 4 1 5 0 0797 2 The loss in electrolytic capacitors is Pross C c ao i Rc 0 487 0 05 0 01W STEP 10 Control Circuit Configuration Figure 24 shows the typical circuit configuration for the RT pin of FLS XS series where the opto coupler transistor is connected to the RT pin to control the switching frequency The minimum switching frequency occurs when the opto coupler transistor is fully tuned off which is given as Le n x 100 kHz min 31 Assuming the saturation voltage of the opto coupler transistor is 0 2V the maximum switching frequency is determined as 5 2kQ k 4 68kQ R x 100 kHz er es 32 min max 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE LVocc VDL nial S S SG PG Figure 24 Typical Circuit Configuration for RT Pin Soft Start To prevent excessive inrush current and overshoot of output voltage
7. larger than the LC series resonant inductor L the magnetizing inductance in an LLC resonant converter is just 3 8 times L which is usually implemented by introducing an air gap in the transformer www fairchildsemi com AN 9729 Figure 2 Half Bridge LLC Resonant Converter An LLC resonant converter has many advantages over a series resonant converter It can regulate the output over wide line and load variations with a relatively small variation of switching frequency It can achieve zero voltage switching ZVS over the entire operating range All essential parasitic elements including junction capacitances of all semiconductor devices and the leakage inductance and magnetizing inductance of the transformer are utilized to achieve soft switching This application note presents design considerations of an LLC resonant half bridge converter employing Fairchild s FLS XS series It includes explanation of the LLC resonant converter operation principles designing the transformer and resonant network and selecting the components The step by step design procedure explained with a design example helps design the LLC resonant converter LLC Resonant Converter and Fundamental Approximation Figure 3 shows a simplified schematic of a half bridge LLC resonant converter where Lm is the magnetizing inductance that acts as a shunt inductor L is the series resonant inductor and C is the resonant capacitor Figure 4 illustrate
8. Actual capacitor selection should be based on the Over Current Protection OCP trip point With the OCP level IOCP the maximum resonant capacitor voltage is obtained as 24 V max I Ve nom in OCP 25 i 2 Leme Ja C Design Example Tc BMS z ei alo 42 nV V er 2 Ep Sn 4 2 fM L L 1 2 22 100 0 9 2 092 ET 2 22 442 99x103 1 12 680x107 0 78A The peak current in the primary side in normal operation is ge V2 I 7 1 1034 OCP level is set to 1 75A with 50 margin on Ic EA Io RMS a m Daf C n V2 0 78 2 2 m 99x10 15x107 max nom Pa ve i 2 i max F max Vin Vo 2 locp ee on 1 75 4 e O 2 2 m 99x10 15x10 A 630V rated low ESR film capacitor is selected for the resonant capacitor STEP 9 Rectifier Network Design When the center tap winding is used in the transformer secondary side the diode voltage stress is twice of the output voltage expressed as Vp 2 ae Vr 26 The RMS value of the current flowing through each rectifier diode is given as I RMS ay 4 0 Meanwhile the ripple current flowing through output capacitor is given as 27 I 28 www fairchildsemi com AN 9729 The voltage ripple of the output capacitor is T AV ae Re 29 where Rc is the effective series resistance ESR of the output capacitor and the power dissipation is the output capacitor is Io Re Lig Co
9. March April 1991 pp 386 395 6 R Oruganti J Yang and F C Lee State Plane Analysis of Parallel Resonant Converters Proc IEEE PESC 85 1985 APPLICATION NOTE 7 M Emsermann An Approximate Steady State and Small Signal Analysis of the Parallel Resonant Converter Running Above Resonance Proc Power Electronics and Variable Speed Drives 91 1991 pp 9 14 8 Yan Liang Wenduo Liu Bing Lu van Wyk J D Design of integrated passive component for a 1MHz 1kW half bridge LLC resonant converter JAS 2005 pp 2223 2228 9 B Yang F C Lee M Concannon Over current protection methods for LLC resonant converter APEC 2003 pp 605 609 10 Yilei Gu Zhengyu Lu Lijun Hang Zhaoming Qian Guisong Huang Three level LLC series resonant DC DC converter IEEE Transactions on Power Electronics Vol 20 July 2005 pp 781 789 11 Bo Yang Lee F C A J Zhang Guisong Huang LLC resonant converter for front end DC DC conversion APEC 2002 pp 1108 1112 12 Bing Lu Wenduo Liu Yan Liang Fred C Lee Jacobus D Van Wyk Optimal design methodology for LLC Resonant Converter APEC 2006 pp 533 538 This application note written based on Fairchild Semiconductor Application Note AN 4137 Related Datasheets FLS1800XS Half Bridge LLC Resonant Control IC for Lighting FLS2100XS Half Bridge LLC Resonant Control IC for Lighting FAN7346 4 Channel LED Cur
10. e a4 o amp P TT 6 ETLE Figure 29 Example of CC and OVR Feedback Circuit www fairchildsemi com AN 9729 APPLICATION NOTE Design Summary design example EER3543 core with sectional bobbin is Figure 30 and Figure 31 show the final schematic of the i G n l a oa r D a n for the transformer Efficiency at full load is around C D201 Dgoot c R102 Wok 10K T RoE 2N2222 LVec R103 CBooT C201 R202 R201 CS Rsense Figure 30 Final Schematic of Half Bridge LLC Resonant Converter for Single Channel D201 C201 C202 C203 VLED Vprcout 15V 2N2907 D203 2N2222 9VELNVA Rsi9 RS20 RS21 Rso7 RS28 RS24 CS17 RseNse l vLSU GISH LISY 8LSu ZZS NN a i A T CS14 CS15 C812 CS16 Figure 31 Final Schematic of Half Bridge LLC Resonant Converter for Multi Channel 2011 Fairchild Semiconductor Corporation www fairchildsemi com Rev 1 0 0 3 22 11 15 AN 9729 Experimental Verification To show the validity of the design procedure presented in this application note the converter of the design example was built and tested All the circuit components are used as designed in the design example Figure 32 and Figure 33 show the operation waveforms at full load and no load conditions for nominal input voltage As observed the MOSFET drain to source voltage Vps drops to zero by resonance before the MOSFE
11. of a LC series resonant converter is always lt 1 At light load condition the impedance of the load is large compared to the impedance of the resonant network all the input voltage is imposed on the load This makes it difficult to regulate the output at light load Theoretically frequency should be infinite to regulate the output at no load 1 L gt HN Vo L Half Bridge LC Series Resonant Converter n Figure 1 To overcome the limitation of series resonant converters the LLC resonant converter has been proposed The LLC resonant converter is a modified LC series resonant converter implemented by placing a shunt inductor across the transformer primary winding as depicted in Figure 2 When this topology was first presented it did not receive much attention due to the counterintuitive concept that increasing the circulating current in the primary side with a shunt inductor can be beneficial to circuit operation However it can be very effective in improving efficiency for high input voltage applications where the switching loss is more dominant than the conduction loss In most practical designs this shunt inductor is realized using the magnetizing inductance of the transformer The circuit diagram of LLC resonant converter looks much the same as the LC series resonant converter the only difference is the value of the magnetizing inductor While the series resonant converter has a magnetizing inductance
12. of the square wave voltage Vz applied to the half bridge totem pole which allows the MOSFETs to be turned on with zero voltage As shown in Figure 4 the MOSFET turns on while the voltage across the MOSFET is zero by flowing current through the anti parallel diode The rectifier network produces DC voltage by rectifying the AC current with rectifier diodes and a capacitor The rectifier network can be implemented as a full wave bridge or center tapped configuration with capacitive output filter Square Wave Generator Rectifier Network Resonant Network Figure 3 Schematic of Half Bridge LLC Resonant Converter Figure 4 Typical Waveforms of Half Bridge LLC Resonant Converter The filtering action of the resonant network allows use of the fundamental approximation to obtain the voltage gain of the resonant converter which assumes that only the fundamental component of the square wave voltage input to the resonant network contributes to the power transfer to the output Because the rectifier circuit in the secondary side acts as an impedance transformer the equivalent load resistance is different from actual load resistance Figure 5 shows how this equivalent load resistance is derived The www fairchildsemi com AN 9729 primary side circuit is replaced by a sinusoidal current source and a square wave of voltage Vrz appears at the input to the rectifier Since the average of 1s the output current
13. output power This can cause SLP or thermal 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 RIO APPLICATION NOTE stress problems in the other channel OLP function has auto recovery As soon as drain voltage is higher than 0 3V OLP is finished and drain voltage feedback system is restored To sense over current condition the FAN7346 monitors FBx pin voltage If FBx voltage is higher than 1 V for 20us CHx is considered in over current condition After sensing OCP condition individual channel switch is latched off So even if a channel is in OCP condition other channels keep operating Any OCP channel is restarted after UVLO 1s reset Design Example The output voltage Vo is 100V in i design target Vo is determined as R8 7 1 5 1 i RIO Set the upper side feedback resistance R8 as IMQ R10 is determined as 1 5xR8 _ 1 5x1MQ 15 L V 1 5 00 1 5 The output channel current Lep is 250mA in design target Setting the Vapi is above 4V the current sense Rsgnse iS _ determined as VAaDIM _ AV O o O SENSE 10x pp 10x 250mA Choose the sense resistor R29 R30 R31 and R32 is 1 5Q the OCP level is determined as _Voce_to W loc 666mA R 1 5 SENSE Vien o rs WT e e a 02 k 03
14. value results in poor coupling of the transformer and deteriorates the efficiency It 1s typical to set m to be 3 7 which results in a voltage gain of 1 1 1 2 at the resonant frequency fo With the chosen m value the voltage gain for the nominal PFC output voltage is obtained as Mm m f f Vim 1 which would be the minimum gain because the nominal PFC output voltage is the maximum input voltage Vn 14 The maximum voltage gain is given as Design Example The ratio m between L and L is chosen as 5 The minimum and maximum gains are _ obtained as mmn RO _ aS AE Vin tea 5 1 2 O MOS _ Vin min 400 1 19 1 93 y min 364 Gain M Peak Gain Available Maximum Gain 1 23 min APPLICATION NOTE STEP 4 Calculate Equivalent Load Resistance With the transformer turns ratio obtained from Equation 16 the equivalent load resistance is obtained as Design Example a oe z 100 O STEP 5 Design the Resonant Network With m value chosen in STEP 2 read proper Q value from the peak gain curves in Figure 14 that allows enough peak gain Considering the load transient and stable zero voltage switching ZVS operation 10 20 margin should be introduced on the maximum gain when determining the peak gain Once the Q value is determined the resonant parameters are obtained as R 405Q l CS 270 f R 18 l Ln T Raf YC m L m L 20 Design Example
15. 0 of switching period with 1kQ resistor a and 100pF capacitor 2 Considering the output voltage overshoot during transient 10 and the controllability Gf the feedback loop the AEA E A EE A E maximum frequency is set as 140kHz Rmax is determined as STEP 12 Voltage and Current Feedback 4 68KQ Power supplies for LED lighting must be controlled by Rmax Ff x140 5 2KQ_ _ Constant Current CC Mode as well as a Constant Voltage C00 KHz R CV Mode Because the forward voltage drop of LED is varies with the junction temperature and the current also 4 68KQ 00KHex140 52KO 7 8KQ Increases greatly consequently devices can be damaged eS Figure 28 shows an example of a CC and CV Mode 100KAz 6 5KQ feedback circuit for single output LED power supply During normal operation CC Mode is dominant and CV control circuit does not activate as long as the feedback voltage is lower than reference voltage which means that Setting the initial frequency of soft start as 250kHz 2 5 times of the resonant frequency the soft start resistor Rss _ is given as 7 Reo 5 2KO CV control circuit only acts as OVP for abnormal modes SSF _AOKHz 52KQ i 52KO aa A OO 100KHz Rat i 7 Design Example The output voltage Vo is 100V in 52KO 2 2 design target Vo is determined as 250KHZ 40KHz oo i Rev C TOKE 6 5KQ i fo lar ae ee ces i Set the upper side feedback resistance Rpy as 330KQ i
16. 2222S FAIRCHILD R SEMICONDUCTOR AN 9729 www fairchildsemi com LED Application Design Guide Using Half Bridge LLC Resonant Converter for 100W Street Lighting Introduction This application note describes the LED driving system using a half bridge LLC resonant converter for high power LED lighting applications such as outdoor or street lighting Due to the existence of the non isolation DC DC converter to control the LED current and the light intensity the conventional PWM DC DC converter has the problem of low power conversion efficiency The half bridge LLC converter can perform the LED current control and the efficiency can be significantly improved Moreover the cost and the volume of the whole LED driving system can be reduced Consideration of LED Drive LED lighting is rapidly replacing conventional lighting sources like incandescent bulbs fluorescent tubes and halogens because LED lighting reduces energy consumption LED lighting has greater longevity contains no toxic materials and emits no harmful UV rays which are 5 20 times longer than fluorescent tubes and incandescent bulbs All metal halide and fluorescent lamps including CFLs n contain mercury The amount of current through an LED determines the light it emits The LED characteristics determine the forward voltage necessary to achieve the required level of current Due to the variation in LED voltage versus current characteristics controlling onl
17. A A i li i ammar bungi k pe rag Ip 1A div lt 257V F aa m H Time 5ys div b j Vp 100V div Figure 35 Rectifier Diode Voltage and Current Waveforms at Full Load Condition Vo P_CC 2V d iv ar A 000mA 140mA lt Transient Modea CC Mode MERN NN 0 t loan 0 5A div Loan Time 50ms div Figure 36 Soft Start Waveforms ViLep 100V div _ i late LED Restore LED ILED_ OPEN 200mA div Y VDS_NORMAL_LED 500mV div VDSD_OPEN_LED 500mV div Sy m Figure 37 Open LED Protection Operation www fairchildsemi com AN 9729 References 1 Robert L Steigerwald A Comparison of Half bridge resonant converter topologies JEEE Transactions on Power Electronics Vol 3 No 2 April 1988 2 A F Witulski and R W Erickson Design of the series resonant converter for minimum stress IEEE Transactions on Aerosp Electron Syst Vol AES 22 pp 356 363 July 1986 3 R Oruganti J Yang and F C Lee Implementation of Optimal Trajectory Control of Series Resonant Converters Proc IEEE PESC 87 1987 4 V Vorperian and S Cuk A Complete DC Analysis of the Series Resonant Converter Proc IEEE PESC 82 1982 5 Y G Kang A K Upadhyay D L Stephens Analysis and design of a half bridge parallel resonant converter operating above resonance IEEE Transactions on Industry Applications Vol 27
18. From STEP 2 the maximum voltage gain M for the minimum input voltage V is 1 23 With 15 margin a peak gain of 1 41 is required m has been chosen as 5 in STEP 2 and Q is obtained as 0 42 from the peak gain curves in i a ao ail for Vin i Figure 19 By selecting the resonant frequency as 100kHz the _ resonant components are determined as E l l a E ee 2aQ fo Rac 2r 0 42 100x10 405 ymin te S re Vo prc l 1 A fs fo Figure 18 Maximum Gain Minimum Gain STEP 3 Determine the Transformer Turns Ratio n N N With the minimum gain M obtained in STEP 2 the transformer turns ratio is given as y max P in 2V V min 16 drop Design Example assuming V is 0 9V ee N 2Vo V 2 100 0 9 Peseeeseenseeneeeneeseeneeeeeenee seen eeseeeseeesenseeeeeseeeseeeseeeeeeeee eee seeneeeeee seen sere eeeeu seen eeeeeeseeeseeeseeeeeeeeseeeseeeeeeeeeseee sere eeeee seen sere eeseeeseeseeeereeeseeeseeeseeQeeseee eee eeeeeeseee seen eerSeeseu seen eereeeseeeseeesereeeeresesd 112 2 22 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 af C 20 x100x103 9 35x10 Ly m L 13554H 1 7 p 1 6 f ne 1 5 n WAM 14 0 x y 1 3 E LJ tee m 4 0 m 4 5 Q E o nP o m 6 0 750 9 0 m 8 0 m 7 0 1 i 0 2 0 3 0 4 0 5 0 6 0 7 0 8 0 9 1 1 1 Figure 19 Resonant Network Design Using the Peak Gain Attainable Maxim
19. STEP 11 Current Sensing and Protection Rri is determined as FLS XS series senses low side MOSFET drain current as a ee 2 5xRry _ 2 5x330KQ _ 8 46KQ negative voltage as shown in Figure 26 and Figure 27 Vo 2 5 100 2 5 Half wave sensing allows low power dissipation in the sensing resistor while full wave sensing has less switching noise in the sensing signal Typically RC low pass filter is V y sense REF used to filter out the switching noise in the sensing signal 9 R201 g R203 sC201 Voc The output voltage of op amp is given as The RC time constant of the low pass filter should be 1 n ee 1 100 1 20 of the switching period R201 R203 C V cn l V conse ah VREF OC 7 ie i sC201 R201 R203 7 Np Ens Actually the Vense has a negative value and assume all i J E F _ resistors have the same value for simplification age nae ig l Ct Tog okk ete Vrer ci i A j T i i _ The output voltage of the op amp for CC control keeps zero Rass A L gt voltage as long as the sensing voltages are lower than the es a ieee 2 reference voltage T E ieee aaeeeercntenwecatnenr anne ig A r C201 R202 R201 Cufent A Feedback N gt Vos a z C202 R205 LLLLZ a C 4 Control f TE ie F i al gei oa S a Ke Np Ns
20. T is turned on and zero voltage switching is achieved Figure 34 shows the waveforms of the resonant capacitor voltage and primary side current at full load condition The peak values of the resonant capacitor voltage and primary side current are 300V and 1 1A respectively which are well matched with the calculated values in STEP 8 of design procedure section Figure 35 shows the rectifier diode voltage and current waveforms at full load condition Due to the voltage overshoot caused by stray inductance the voltage stress is a little bit higher than the value calculated in STEP 9 Figure 36 shows the output load current and output voltage of op amp waveforms for constant current control when output load is step changed from 140mA to 1000mA at to Figure 37 shows the operation waveform when LED string is opened and restored condition iP 1 pidiv AVA lbs 1 Ndiv es ma mai wa Vos 200vid iv l Time 5us div Figure 32 Operation Waveforms at Full Load Condition lp 2A div lbs 1 Aldiv vce cien danaeendinde Vos 200V d iv _ _ 1 a Time 5us div Figure 33 Operation Waveforms at No Load Condition 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE alibi Ver 200V div a Vps 200V div eA AAAA V V AN EN Figure 34 Resonant Capacitor Voltage and Primary Side Current Waveforms at Full Load Condition Ip 1A div A
21. aximum gain varies with QO for different m values It appears that higher peak gain can be obtained by reducing m or Q values With a given resonant frequency f and Q value decreasing m means reducing the magnetizing inductance which results in increased circulating current There is a trade off between the available gain range and conduction loss 2 2 p 2 1 2 L 1 9 7 1 8 7 lai 1 6 7 m 2 25 m 2 5 1 3 F m 3 0 m 3 5 1 2 f m 4 0 m 4 5 m 5 0 m 6 0 m 9 0 m 8 0 m 7 0 Peak Gain D Jel r Figure 14 Peak Gain Attainable Maximum Gain vs Q for Different m Values www fairchildsemi com AN 9729 Features of FLS XS Series FLS XS_ series is an integrated Pulse Frequency Modulation PFM controller and MOSFETs specifically designed for Zero Voltage Switching ZVS half bridge converters with minimal external components The internal controller includes an under voltage lockout optimized high side low side gate driver temperature compensated precise current controlled oscillator and self protection circuitry Compared with discrete MOSFET and PWM controller solutions FLS XS series can reduce total cost component count size and weight while simultaneously increasing efficiency productivity and system reliability 1 2345 6 7 8 9 10 VoL Rr SG LVec AR CS PG HVcc Figure 15 Package Diagram t T LVce Good
22. btain the desired L value For a sectional bobbin the number of turns and winding configuration are the major factors determining the value of L while the gap length of the core does not affect L Figure 20 Flux Density Swing much L can be controlled by adjusting the gap length Choose the proper number of turns for the secondary side Table 2 shows measured L and L values with different that results in primary side turns larger than N as gap lengths A gap length of 0 05mm obtains values for L and L closest to the designed parameters Sion gt N 22 Design Example EER3542 core A 107mm is selected for the transformer From the gain curve of Figure 21 the minimum switching frequency is obtained as 2 70KHz The minimum primary side turns of the transformer is given as ymin Vo e ar 2 2AN Figure 22 Sectional Bobbin a 00205 2 2x80x10 0 4 1 11 107x10 Choose N so that the resultant N is larger than Nee N n N 2 02x13 29 lt i ia Table 2 Measured L and L with Different Gap Lengths ON n N 2 22x15 33 gt N 2 295uH 123uH N n N 2 22x16 36 gt N 9434H 1224H ee _ _ min EE 19H 15 2011 Fairchild Semiconductor Corporation www fairchildsemi com Rev 1 0 0 3 22 11 10 AN 9729 Design Example Final Resonant Network Design Even though the integrated transformer approach in LLC 2 resonant converter design can implement
23. during startup increase the voltage gain of the resonant converter progressively Since the voltage gain of the resonant converter is reversely proportional to the switching frequency soft start is implemented by sweeping down the switching frequency from an initial high frequency f until the output voltage is established as illustrated in Figure 25 The soft start circuit is made by connecting RC series network on the RT pin as shown in Figure 24 FLS XS series also has an internal soft start for 3ms to reduce the current overshoot during the initial cycles which adds 40KHz to the initial frequency of the external soft start circuit as shown in Figure 25 The actual initial frequency of the soft start is given as pS eZ n 5 2kQ Rmin SS It is typical to set the initial frequency of soft start f 2 3 times of the resonant frequency f x 100 40 kHz 33 ISS The soft start time is determined by the RC time constant Tx 3 4 times of Ry C5 34 fs Control Loop _ Take Over Time Figure 25 Frequency Sweep of the Soft Start www fairchildsemi com AN 9729 APPLICATION NOTE Design Example The minimum frequency is 80kHz in Design Example Since the OCP level is determined as STEP 6 Rmin is determined as _ 1 75A in STEP 8 and the OCP threshold voltage is 0 6V a 100KHz sensing resistor of 0 33Q is used The RC time constant is i Rmin l x 5 2KO 6 5KO set to 100ns 1 10
24. e capacitive region is that the output voltage becomes out of control since the slope of the gain is reversed The minimum switching frequency should be limited above the peak gain frequency www fairchildsemi com AN 9729 M i Capacitive ae Inductive Region Peaks Region Reverse Recovery ZVS Figure 12 Operation Waveforms for Capacitive and Inductive Regions The available input voltage range of the LLC resonant converter is determined by the peak voltage gain Thus the resonant network should be designed so that the gain curve has an enough peak gain to cover the input voltage range However ZVS condition is lost below the peak gain point as depicted in Figure 12 Therefore some margin 1s required when determining the maximum gain to guarantee stable ZVS operation during the load transient and startup Typically 10 20 of the maximum gain is used as a margin as shown in Figure 13 Gain M a Peak Gain 10 20 of M Maximum Operation aaa aa Gain Min fo fs Figure 13 Determining the Maximum Gain 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE Even though the peak gain at a given condition can be obtained using the gain in Equation 6 it is difficult to express the peak gain in explicit form To simplify the analysis and design the peak gains are obtained using simulation tools and depicted in Figure 14 which shows how the peak gain attainable m
25. fficiency the maximum input power is given as f 11 f Input Voltage Range Vi and Va e The maximum input voltage would be the nominal PFC output voltage as y in min V0 PFC 12 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 2 min V z Vo prc Even though the input voltage is regulated as constant by PFC pre regulator it drops during the hold up time The minimum input voltage considering the hold up time requirement is given as min 2 in Vi Vo pre 13 Chr where Vo prc is the nominal PFC output voltage Tyy 1s a hold up time and Cp is the DC link bulk capacitor f Design Example Assuming the efficiency is 92 21 oa Ep 0 92 l VP V pro 400V 2 2E T HU Cpr 3 E hog 2 109 al E 240x107 n STEP 2 Determine Maximum and Minimum Voltage Gains of the Resonant Network As discussed in the previous section it is typical to operate the LLC resonant converter around the resonant frequency fo to minimize switching frequency variation Since the input of the LLC resonant converter is supplied from PFC output voltage the converter should be designed to operate at f for the nominal PFC output voltage www fairchildsemi com AN 9729 As observed in Equation 10 the gain at f is a function of m m L L The gain at f is determined by choosing that value of m While a higher peak gain can be obtained with a small m value too small m
26. ion loss than the below resonance operation It can show better efficiency for low output voltage applications such as Liquid Crystal Display LCD TV or laptop adaptor where Schottky diodes are available for the secondary side_ rectifiers and reverse recovery problems are insignificant However operation above the resonant frequency may cause too much frequency increase at light load condition Above frequency operation requires frequency skipping to prevent too much increase of the switching frequency 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE ve ene SOC 6 Seeen Above Resonance fs gt fo Below Resonance fs lt fo fo fs Figure 10 Operation Modes According to the Operation Frequency Figure 11 Waveforms of Each Operation Mode Required Maximum Gain and Peak Gain Above the peak gain frequency the input impedance of the resonant network is inductive and the input current of the resonant network lags the voltage applied to the resonant network Va This permits the MOSFETs to turn on with zero voltage ZVS as illustrated in Figure 12 Meanwhile the input impedance of the resonant network becomes capacitive and J leads V4 below the peak gain frequency When operating in capacitive region the MOSFET body diode is reverse recovered during the switching transition which results in severe noise Another problem of entering th
27. ith a given transformer In an actual transformer L and L can be measured in the primary side with the secondary side winding open circuited and short circuited respectively In Figure 9 notice that a virtual gain My is introduced which is caused by the secondary side leakage inductance By adjusting the gain equation of Equation 6 using the modified equivalent circuit of Figure 9 the gain equation for integrated transformer is obtained by o 2 m 2m va Gye i M no E n E Sn mn O o Cmm i E eC ao O 0 where 9 l eo _4 T Me i r a l 2 eaa A a ans The gain at the resonant frequency o is fixed regardless of the load variation which is given as m ato a 10 The gain at the resonant frequency is unity when using individual core for series inductor as shown in Equation 7 However when implementing the magnetic components with integrated transformer the gain at the resonant frequency is larger than unity due to the virtual gain caused by the leakage inductance in the transformer secondary side The gain of Equation 9 is plotted in Figure 10 for different Q values with m 3 f 100kHz and f 57kHz As observed in Figure 9 the LLC resonant converter shows gain characteristics almost independent of the load when the switching frequency is around the resonant frequency fo www fairchildsemi com AN 9729
28. n LED Protection OLP Short LED Protection SLP and Over Current Protection OCP which increase system reliability are applied in individual string protection method To sense a short LED condition the FAN7346 senses drain voltage level If LEDs are shorted the LED forward voltage is lower than other LED strings so its drain voltage of external balance switch is higher than other drain voltage The SLP condition detection threshold voltage can be programmed by SLPR voltage The internal short LED protection reference is determined as Vsrp_tH 10xVsrpr 36 Minimum SLP threshold voltage is 0V and maximum SLP threshold voltage is 45V If any string is in SLP condition SLP string is turned off and other string is operated normally If the sensed drain voltage CHx voltage is higher than the programmed threshold voltage for 20us CHx goes to short LED protection As soon as encountering SLP the corresponding channel is forced off To sense an open LED condition the FAN7346 senses drain voltage level If LED string is opened its drain voltage of external balance switch is grounded so the FAN7346 detects the open LED condition The detection threshold voltage is 0 3V If CHx voltage is lower than 0 3V for 20us its drain voltage feedback is pulled up to 5V This means the opened LED string is eliminated from drain feedback loop Without OLP minimum drain voltage is OV so drain voltage feedback forces the FB signal to increase
29. rent Balance Control IC DISCLAIMER FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY FUNCTION OR DESIGN FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS NOR THE RIGHTS OF OTHERS LIFE SUPPORT POLICY FAIRCHILD S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR CORPORATION As used herein 1 Life support devices or systems are devices or systems which a are intended for surgical implant into the body or b support or sustain life or c whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in significant injury to the user 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 2 A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness www fairchildsemi com
30. s the typical waveforms of the LLC resonant converter It is assumed that the operation frequency is same as the resonance frequency determined by the resonance between L and C Since the magnetizing inductor is relatively small a considerable amount of magnetizing current Im exists which freewheels in the primary side without being involved in the power transfer The primary side current is sum of the magnetizing current and the secondary side current referred to the primary In general the LLC resonant topology consists of three stages shown in Figure 3 square wave generator resonant network and rectifier network The square wave generator produces a square wave voltage V4 by driving switches Q and Q alternately with 50 duty cycle for each switch A small dead time is usually introduced between the consecutive transitions The square wave generator stage can be built as a full bridge or half bridge type The resonant network consists of a capacitor leakage inductances and the magnetizing inductance of the transformer The resonant network filters the higher harmonic currents Essentially only sinusoidal current is allowed to flow through the resonant 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE network even though a square wave voltage is applied to the resonant network The current lags the voltage applied to the resonant network that is the fundamental component
31. shows the gain is unity at resonant frequency w regardless of the load variation which is given as 2n V _ m 1 a l atwa a 7 y wo O The gain of Equation 6 is plotted in Figure 7 for different Q values with m 3 f 100kHz and f 57kHz As observed in Figure 7 the LLC resonant converter shows gain characteristics that are almost independent of the load when the switching frequency is around the resonant frequency f This is a distinct advantage of LLC type resonant converter over the conventional series resonant converter Therefore it is natural to operate the converter around the resonant frequency to minimize the switching frequency variation The operating range of the LLC resonant converter is limited by the peak gain attainable maximum gain which is indicated with in Figure 7 Note that the peak voltage gain does not occur at fe or fp The peak gain frequency where the peak gain is obtained exists between www fairchildsemi com AN 9729 J and fo as shown in Figure 7 As Q decreases as load decreases the peak gain frequency moves to f and higher peak gain is obtained Meanwhile as Q increases as load increases the peak gain frequency moves to f and the peak gain drops the full load condition should be worst case for the resonant network design 1 zaJ a 1 0 Q 0 75 2 A Q 0 50 N Q 0 25 N ne
32. t S 80 90 100 110 120 130 140 Frequency kHz Figure 7 Typical Gain Curves of LLC Resonant Converter m 3 Consideration for Integrated Transformer For practical design it is common to implement the magnetic components series inductor and shunt inductor using an integrated transformer where the leakage inductance is used as a series inductor while the magnetizing inductor is used as a shunt inductor When building the magnetizing components in this way the equivalent circuit in Figure 6 should be modified as shown in Figure 8 because leakage inductance exists not only in the primary side but also in the secondary side Not considering the leakage inductance in the transformer secondary side generally results in an ineffective design L L L ee E lkp Lip La 11 L kp Ideal Transformer i Figure 8 Modified Equivalent Circuit to Accommodate the Secondary Side Leakage Inductance 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE In Figure 8 the effective series inductor L and shunt inductor L L are obtained by assuming 1 Liks Likp and referring the secondary side leakage inductance to the primary side as L L L lkp L Ly L Mn Le Ly laL lkp 8 lkp When handling an actual transformer equivalent circuit with L and L is preferred since these values can be measured w
33. the magnetic components in a single core and save one magnetic i component the value of L is not easy to control in real i transformer design Resonant network design sometimes i requires iteration with a resultant L value after the i transformer is built The resonant capacitor value is also changed since it should be selected among off the shelf network design is summarized in Table 3 and the new gain curves are shown capacitors The final resonant in Figure 23 Table 3 Final Resonant Network Design Parameters 100 load 80 load 60 load 40 load 20 load f normal Gain 40 50 60 70 80 90 100 110 120 130 140 Frequency KHz Figure 23 Gain Curve of the Final Resonant Network Design STEP 8 Select the Resonant Capacitor When choosing the resonant capacitor the current rating should be considered because a considerable amount of current flows through the capacitor The RMS current through the resonant capacitor is given as Lz nV V I p te _p 23 E Nan N2 f M L L 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 APPLICATION NOTE The nominal voltage of the resonant capacitor in normal operation is given as y rom x Va fe V2 I 2 LRC However the resonant capacitor voltage increases much higher at overload condition or load transient
34. um Gain Curve for m 5 www fairchildsemi com AN 9729 APPLICATION NOTE STEP 6 Design the Transformer EF NK 100 load 80 load The worst case for the transformer design is the minimum 18 l Ta oye a loa switching frequency condition which occurs at the l 20 load minimum input voltage and full load condition To obtain 16 AI the minimum switching frequency plot the gain curve l using gain Equation 9 and read the minimum switching 3 frequency The minimum number of turns for the transformer primary side is obtained as es Mm N min _ nV V gt 21 Mein p 7 min 1 0 2h M AB A where A is the cross sectional area of the transformer re core in m and AB is the maximum flux density swing in Tesla as shown in Figure 20 If there is no reference m data use AB 0 3 0 4 T 2 40 50 6 7 8 90 100 110 120 130 140 Frequency KHz ee 3 H n Vot Ve Mv Figure 21 Gain Curve STEP 7 Transformer Construction Parameters L and L of the transformer were determined in a see faH A VotVEVMy STEP 5 L and L can be measured in the primary side with the secondary side winding open circuited and short circuited respectively Since LLC converter design requires a relatively large L a sectional bobbin is typically used as shown in Figure 22 to o
35. y the voltage across the LED leads to variability in light output Therefore most LED drivers use current regulation to support brightness control Brightness can be controlled directly by changing the LED current Consideration of LLC Resonant Converter The attempt to obtain ever increasing power density of switched mode power supplies has been limited by the size of passive components Operation at higher frequencies considerably reduces the size of passive components such as transformers and filters however switching losses have been an obstacle to high frequency operation To reduce switching losses and allow high frequency operation resonant switching techniques have been developed These techniques process power in a sinusoidal manner and the switching devices are softly commutated Therefore the switching losses and noise can be dramatically reduced 2011 Fairchild Semiconductor Corporation Rev 1 0 0 3 22 11 Among various kinds of resonant converters the simplest and most popular is the LC series resonant converter where the rectifier load network is placed in series with the L C resonant network as depicted in Figure 1 In this configuration the resonant network and the load act as a voltage divider By changing the frequency of driving voltage Vq the impedance of the resonant network changes The input voltage is split between this impedance and the reflected load Since it is a voltage divider the DC gain
Download Pdf Manuals
Related Search