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LINEAR TECHNOLOGY LT1506 Manual(1)

1. 5V Lead Temperature Soldering 10 sec 300 C PACKAGE ORDER INFORMATION FRONT VIEW ri y Mri iud pee E IN SW gt LTISO6CR 3 3 TBOR Ds DT sDGCS833 GND sw 3 FB OR 5 3 W T45060R SYNC SENSE 1 SON 171506198 R PACKAGE LT1506CR 3 3 SYNC i LT15061S8 3 3 7 LEAD PLASTIC DD PAK LT1506IR T Tumax 125 Oja 30 C W LT1506IR 3 3 8 PART MARKING WITH PACKAGE SOLDERED TO 0 5 SQUARE INCH 64 80 C W COPPER AREA OVER BACKSIDE GROUND PLANE OR LT1506IR SYNC WITH FUSED GND GROUND PIN ome TS LT15061R 3 3 SYNC MEO 1506 1508 gt 150633 506133 Default is the adjustable output voltage device with FB pin and shutdown function Option 3 3 replaces FB with SENSE pin for fixed 3 3V output applications SYNC replaces SHDN with SYNC pin for applications requiring synchronization Consult factory for Military grade parts ELECTRICAL CHARACTERISTICS Ty 25 C Viy BV Ve 1 5V Boost Viy 5V switch open unless otherwise noted PARAMETER CONDITIONS MIN TYP MAX UNITS Feedback Voltage Adjustable 2 39 242 2 45 V All Conditions e 236 2 48 V Sense Voltage Fixed 3 3V 3 25 3 3 3 35 V All Conditions e 323 3 37 V SENSE Pin Resistance 4 6 6 9 5 kQ Reference Voltage Line Regulation 4 3V x Vy lt 15V 0 0 0 03 V Feedback Input Bias Current e 0 5 2 Error Amplifier Voltage Gain Notes 2 8 200 400 Error Amplifier Transconductance Al Vc 10pA
2. 1506 G08 Feedback Pin Voltage 2 430 V FEEDBACK VOLTAGE 3000 2500 INPUT SUPPLY CURRENT uA na D Ds a 2 420 2 415 0 25 50 TEMPERATURE C 1506 G03 Shutdown Supply Current 25 VSHDN OV N 5 10 INPUT VOLTAGE V 15 1506 G06 Error Amplifier Transconductance 930 3SVHd Rou 200k Vep 2x10 3 ERROR AMPLIFIER EQUIVA Vl TI 90Q LIIN I 1k LENT CIRCUIT 50 10k 100k 10M FREQUENCY Hz 1M 1506 G09 LT 1506 TYPICAL PERFORMANCE CHARACTERISTICS Frequency Foldback 500 400 WITCHING FREQUENCY 300 200 100 FEEDBACK PIN CURRENT SWITCHING FREQUENCY kHz OR CURRENT uA 0 0 5 1 0 1 5 2 0 2 5 FEEDBACK PIN VOLTAGE V 1506 G10 Maximum Load Current at Vout 5V 4A L 10uH 42 4 0 L 5uH 38 3 6 L 3uH 3 34 e 3 2 pas 3 0 L 1 8uH 2 8 2 6 5 7 9 11 13 15 INPUT VOLTAGE V 1506 G17 Current Limit Foldback FOLDBACK CHARACTERISTICS POSSIBLE UNDESIRED CURRENT STABLE POINT FOR SOURCE CURRENT SOURCE LOAD LOAD oc ed ce tas 0 20 40 60 80 100 OUTPUT VOLTAGE 1506 G18 Ko
3. CAD LT1506 TECHNOLOGY A 5A 500kHz Step Down switching Regulator FEATURES DESCRIPTION Constant 500kHz Switching Frequency The LT 1506 is a500kHz monolithic buck mode switching Easily Synchronizable regulator functionally identical to the LT1374 but optimized m Operates with Input as Low as 4V for lower input voltage applications It will operate over a Uses All Surface Mount Components AV to 15V input range compared with 5 5V to 25V for the Inductor Size Reduced to 1 8uH LT1374 A 4 5A switch is included on the die along with all m Saturating Switch Design 0 07Q the necessary oscillator control and logic circuitry High Shutdown Current 200A switching frequency allows a considerable reduction in the m Cycle by Cycle Current Limiting size of external components The topology is current mode for fast transient response and good loop stability Both APPLICATIONS fixed output voltage and adjustable parts are available Aspecial high speed bipolar process and new design tech Portable Computers niques achieve high efficiency at high switching frequency Battery Powered Systems Efficiency is maintained over a wide output current range Battery Charger by keeping quiescent supply current to 4mA and by utiliz m Distributed Power ing a supply boost capacitor to saturate the power switch The LT1506 fits into standard 7 pin DD and fused lead 50 8 packages Full cycle by cycle short circuit protection and thermal shutdown are provide
4. Figure 8 Discontinuous Mode Ringing mitigates this problem but negative voltages over 1V lasting longer than 10ns should be avoided Note that 100MHz oscilloscopes are barely fast enough to see the details of the falling edge overshoot in Figure 7 A second much lower frequency ringing is seen during switch off time if load current is low enough to allow the inductor current to fall to zero during part of the switch off time See Figure 8 Switch and diode capacitance reso nate with the inductor to form damped ringing at 1MHz to 10 MHz This ringing is not harmful to the regulator and it has not been shown to contribute significantly to EMI Any attempt to damp it with a resistive snubber will degrade efficiency INPUT BYPASSING AND VOLTAGE RANGE Input Bypass Capacitor Step down converters draw current from the input supply in pulses The average height of these pulses is equal to load current and the duty cycle is equal to Voy7 Vjy Rise and fall time of the current is very fast A local bypass capacitor across the input supply is necessary to ensure proper operation of the regulator and minimize the ripple current fed back into the input supply The capacitor also forces switching current to flow in a tight local loop minimizing EMI Do not cheat on the ripple current rating of the Input bypass capacitor but also don t get hung up on the value in microfarads The input capacitor is intended to absorb all the switching curre
5. determines output ripple voltage At 500kHz any polarized capacitor is essentially resistive To get low ESR takes volume so physically smaller capacitors have high ESR The ESR range for typical LT1506 applications is 0 050 to 0 20 A typical output capacitor is an AVX type TPS 100uF at 10V with a guaranteed ESR less than 0 10 This is a D size surface mount solid tantalum capacitor TPS capacitors are specially constructed and tested for low ESR so they give the lowest ESR for a given volume The value in microfarads is not particularly critical and values from 22uF to greater than 500pF work well but you cannot cheat mother nature on ESR If you find a tiny 22uF solid tantalum capacitor it will have high ESR and output ripple voltage will be terrible Table 3 shows some typical solid tantalum surface mount capacitors Table 3 Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E Case Size ESR Max O Ripple Current A AVX TPS Sprague 593D 0 1 to 0 3 0 7 to 1 1 AVX TAJ 0 7 to 0 9 0 4 D Case Size AVX TPS Sprague 593D 0 1 to 0 3 0 7 to 1 1 C Case Size AVX TPS 0 2 typ 0 5 typ Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents This is historically true and type TPS capacitors are specially tested for surge capability but surge ruggedness is not a critical issue with the output capacitor Solid tantalum capaci
6. divider To calculate gain and transconductance refer to SENSE pin on fixed voltage parts Divide values shown by the ratio Vour 2 42 LT 1506 TYPICAL PERFORMANCE CHARACTERISTICS Minimum Input Voltage with 3 3V Output 47 4 5 A wo Ez um wo INPUT VOLTAGE V gt Ny ro 2 co eo 10 100 LOAD CURRENT mA 1000 1506 G12 Shutdown Pin Bias Current 500 AT 0 37V SHUTDOWN THRESHOLD AFTER SHUTDOWN CURRENT DROPS TO A FEW uA 400 300 AT 2 38V LOCKOUT THRESHOLD 0 50 25 0 25 50 100 125 TEMPERATURE C 75 1506 G04 Shutdown Supply Current 70 Vin 10V INPUT SUPPLY CURRENT uA 0 1 0 2 0 3 SHUTDOWN VOLTAGE V 0 4 1506 G07 e Mh TRANSCONDUCTANCE Switch Peak Current Limit 6 5 6 0 TYPICAL z 5 5 cc 5 50 x MINIMUM 5 45 x E 40 z4 wn 3 5 3 0 0 20 40 60 80 100 DUTY CYCLE 1506 G02 Lockout and Shutdown Thresholds 2 40 LOCKOUT 2 36 oO 2 32 gt 2 25 a 08 S START UP 8 04 o SHUTDOWN 50 25 0 25 50 75 100 125 JUNCTION TEMPERATURE C 1506 G05 Error Amplifier Transconductance 2500 2000 1500 1000 500 0 50 25 0 25 50 75 100 125 JUNCTION TEMPERATURE C
7. increased by x3 Tolerances in the reference voltages result in small offset currents to flow between the Vc pins The overall effect is that the loop regulates the output at a voltage between the minimum and maximum reference of the devices used Switch current matching between devices will be typically better than 300mA The negative temperature coefficient of the Vc to switch currenttranscon ductance prevents current hogging Acommon Vc voltage forces each LT1506 to operate at the same switch current not duty cycle Each device operates atthe duty cycle defined by its respective input voltage In Figure 15 the input could be split and each device oper ated at a different voltage The common Vo ensures loading is shared between inputs COUNTER D2 1N914 1 8MHz LT1506 SYNC Vc SYNC SW GND Viy BOOST FB LI LI INPUT L1 D03316P 682 LT1506 SYNC Vc SYNC SW GND Viy BOOST FB Synchronized Ripple Currents A ring counter generates three synchronization signals at 600kHz 33 duty cycle phased 120 apart The sync input will operate over a wide range of duty cycles so no further pulse conditioning is needed Each device s maxi mum input ripple current is a 4A square wave at 600kHz When synchronously added together the ripple remains at 4A but frequency increases to 1 8MHz Likewise the output ripple current is a 1 8MHz triangular waveform with maximum amplitude of 350mA at 10V Viy Interest ingly at 7 6V a
8. off to 90 and staying there The overall loop has a gain of 74dB at low frequency rolling off to unity gain at 100kHz Phase shows atwo pole characteristic until the ESR ofthe output capacitor brings it back above 10kHz Phase margin is about 60 at unity gain Analog experts will note that around 4 4kHz phase dips very close to the zero phase margin line This is typical of switching regulators especially those that operate over a wide range of loads This region of low phase is not a problem as long as it does not occur near unity gain In practice the variability of output capacitor ESR tends to dominate all other effects with respect to loop response Variations in ESR will cause unity gain to move around but at the same time phase moves with it so that adequate phase margin is maintained over a very wide range of ESR gt 3 1 930 3SvHd 1500 L Rout Cour Vre x10 3 200k 12pF 50 1000 ERROR AMPLIFIER EQUIVALENT CIRCUIT H H 0 HT RLoap 50Q 500 50 100 1k 10k 100k 1M 10M FREQUENCY Hz 1506 F11 Figure 11 Error Amplifier Gain and Phase 80 200 Pr GAIN 60 150 EN o S 40 100 amp z c Sr ae x a PHASE amp 20 50 c E Vin 10V 0 F Vour 5V lout 2A 0 Cour 100uF 10V AVX TPS 1 5nF Rc 0 1 10uH 20 50 10 100 4k
9. turns on regulating switch current via the Vc pin to maintain a constant dv dt at the output Output rise time is controlled by the current through Css defined by R4 and Q1 s Once the output is in regulation Q1 turns off and thecircuitoperates normally R3 is transient protection for the base of Q1 R4 Css Vour Using the values shown Figure 16 47 10 15 1079 5 0 7 Therampis linear and rise times in the order of 100ms are possible Since the circuit is voltage controlled the ramp rate is unaffected by load characteristics and maximum RiseTime RiseTime 5ms OUTPUT 5V 4A Figure 16 Buck Converter with Adjustable Soft Start output current is unchanged Variants of this circuit can be used for sequencing multiple regulator outputs Dual Output SEPIC Converter The circuit in Figure 17 generates both positive and negative 5V outputs with a single piece of magnetics The two inductors shown are actually just two windings on a standard B H Electronics inductor The topology for the 5V output is a standard buck converter The 5V topology would be a simple flyback winding coupled to the buck converter if C4 were not present C4 creates a SEPIC Single Ended Primary Inductance Converter topology whicn improves regulation and reduces ripple current in L1 Without C4 the voltage swing on L1B compared to L1A would vary due to relative loading and coupling losses C4 provides low
10. 10k 100k 1M FREQUENCY Hz 1505 F12 Figure 12 Overall Loop Characteristics What About a Resistor in the Compensation Network Itis common practice in switching regulator design to add a zero to the error amplifier compensation to increase loop phase margin This zero is created in the external network in the form of a resistor Rc in series with the compensation capacitor Increasing the size of this resis tor generally creates better and better loop stability but there are two limitations on its value First the combina tion of output capacitor ESR and a large value for Rc may cause loop gain to stop rolling off altogether creating a gain margin problem An approximate formula for Rc where gain margin falls to zero is Vour 19 LT 1506 APPLICATIONS INFORMATION Gyp Transconductance of power stage 5 3A V Gma Error amplifier transconductance 2 1073 ESR Output capacitor ESR 2 42 Reference voltage With 5V and ESR 0 030 a value of 6 5k for Rc would yield zero gain margin so this represents an upper limit There is a second limitation however which has nothing to do with theoretical small signal dynamics This resistor sets high frequency gain of the error amplifier including the gain at the switching frequency If switching frequency gain is high enough output ripple voltage will appear at the Vc pin with enough amplitude to muck up proper operation of the regulator In the margina
11. 87W Thermal resistance for LT1506 package is influenced by the presence of internal or backside planes With a full plane under the SO package thermal resistance will be about 80 C W No plane will increase resistance to about 120 C W To calculate die temperature use the proper thermal resistance number for the desired package and add in worst case ambient temperature Ty TA Pror With the 50 8 package Oja 80 C W at an ambient temperature of 50 C T 50 80 0 87 120 C Die temperature is highest at low input voltage so use lowest continuous input operating voltage for thermal calculations FREQUENCY COMPENSATION Loop frequency compensation of switching regulators can be a rather complicated problem because the reactive components used to achieve high efficiency also introduce multiple poles into the feedback loop The inductor and output capacitor on a conventional step down converter actually form a resonant tank circuit that can exhibit peaking and a rapid 180 phase shift at the resonant frequency By contrast the L T1506 uses a cur rent mode architecture to help alleviate phase shift cre ated by the inductor The basic connections are shown in Figure 9 Figure 10 shows a Bode plot of the phase and gain ofthe power section of the LT1506 measured from the Vc pin to the output Gain is set by the 5 3A V transconduc tance of the LT1506 power section and the effective complex impedance from outp
12. AND UNDERVOLTAGE LOCKOUT Figure 4 shows how to add undervoltage lockout UVLO tothe LT1506 Typically ULVO is used in situations where the input supply is current limited or has a relatively high source resistance A switching regulator draws constant power from the source so source current increases as source voltage drops This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions ULVO prevents the regulator from operating at source voltages where these problems might occur Threshold voltage for lockout is about 2 38V slightly less than the internal 2 42V reference voltage A 3 5pA bias current flows out of the pin at threshold This internally generated current is used to force a default high state on the shutdown pin if the pin is left open When low shut down current is not an issue the error due to this current can be minimized by making 10k or less If shutdown current is an issue can be raised to 100k but the error due to initial bias current and changes with temperature should be considered Rio 10k to 100k 25k suggested Rio Vin 2 38V 2 38V 3 5 Vin Minimum input voltage 13 LT 1506 APPLICATIONS INFORMATION LT1506 OUTPUT TOTAL SHUTDOWN 1506 F04 Figure 4 Undervoltage Lockout Keep the connections from the resistors to the shutdown pin short and make sure that i
13. FREQUENCY Vin CIRCULATING PATH AC cdd 1506 Fos Figure 6 High Speed Switching Path 15 LT 1506 APPLICATIONS INFORMATION PARASITIC RESONANCE Resonance or ringing may sometimes be seen on the switch node see Figure 7 Very high frequency ringing following switch rise time is caused by switch diode input capacitor lead inductance and diode capacitance Schot tky diodes have very high Q junction capacitance that can ring for many cycles when excited at high frequency If total lead length for the input capacitor diode and switch path is 1 inch the inductance will be approximately 25nH At switch off this will produce a spike across the NPN Output device in addition to the input voltage At higher currents this spike can be in the order of 10V to 20V or higher with a poor layout potentially exceeding the abso lute max switch voltage The path around switch catch diode and input capacitor must be kept as short as possible to ensure reliable operation When looking at this 100MHz oscilloscope must be used and waveforms should be observed on the leads of the package This switch off spike will also cause the SW node to go below ground The LT1506 has special circuitry inside which RISE AND FALL WAVEFORMS ARE SUPERIMPOSED PULSE WIDTH IS NOT 120ns 20ns DIV 1375 76 FO7 Figure 7 Switch Node Resonance owe SWITCH NODE VOLTAGE INDUCTOR 0 5us DIV 1375 76 F08
14. HIFT CIRCUIT DO NOT FORWARD BIAS FB LOCKOUT COMPARATOR ae ERROR C AMPLIFIER 2 38V Om 2000uMho 2 42V _ GND 1506 BD Figure 1 Block Diagram LT 1506 APPLICATIONS INFORMATION FEEDBACK PIN FUNCTIONS The feedback FB pin on the LT1506 is used to set output voltage and provide several overload protection features The first part of this section deals with selecting resistors to set output voltage and the remaining part talks about foldback frequency and current limiting created by the FB pin Please read both parts before committing to a final design The fixed 3 3V LT1506 3 3 has internal divider resistors and the FB pin is renamed SENSE connected directly to the output The suggested value for the output divider resistor see Figure 2 from FB to ground R2 is 5k or less and a formula for R1 is shown below The output voltage error caused by ignoring the input bias current on the FB pin is less than 0 25 with R2 5k Please read the following if divider resistors are increased above the suggested values R2 Vour 2 42 2 42 LT1506 TO FREQUENCY OUTPUT SHIFTING ERROR AMPLIFIER Figure 2 Frequency and Current Limit Foldback More Than Just Voltage Feedback The feedback pin is used for more than just output voltage sensing It also reduces switching frequency and current limit when output voltage is very low see the Frequency Foldback graph in Typical Performance Characte
15. LT1076 Step Down Switching Regulators 40V Input 100kHz 5A and 2A LTC 1148 High Efficiency Synchronous Step Down Switching Regulator External FET Switches LTC1149 High Efficiency Synchronous Step Down Switching Regulator External FET Switches LTC1174 High Efficiency Step Down and Inverting DC DC Converter 0 5A 150kHz Burst Mode Operation LT1176 Step Down Switching Regulator PDIP LT1076 LT1370 High Efficiency DC DC Converter 42V 6A 500kHz Switch LT1371 High Efficiency DC DC Converter 35V 3A 500kHz Switch LT1372 LT1377 500kHz and 1MHz High Efficiency 1 5A Switching Regulators Boost Topology LT1374 High Efficiency Step Down Switching Regulator 25V 4 5A 500kHz Switch LT1435 LT1436 High Efficiency Step Down Converter External Switches Low Noise Burst Mode is a trademark of Linear echnology Corporation Linear Technology Corporation 1630 McCarthy Blvd Milpitas CA 95035 7417 408 432 1900 FAX 408 434 0507 www linear tech com 1506f LT TP 1198 4K PRINTED IN USA TECHNOLOGY LINEAR TECHNOLOGY CORPORATION 1998
16. Note 8 1500 2000 2700 uMho e 1000 3100 uMho Vc Pin to Switch Current Transconductance 5 3 A V Error Amplifier Source Current Veg 2 1V or Vsense 2 9V e 140 225 320 uA Error Amplifier Sink Current Vrg 2 7V or 3 7V e 140 225 320 pA Vc Pin Switching Threshold Duty Cycle 0 0 9 V Vc Pin High Clamp 2 1 V Switch Current Limit Vc Open Veg 2 1V or Vsense 2 9V DC lt 50 e 4 5 6 8 5 A Slope Compensation DC 80 0 8 A 2 LT 1506 ELECTRICAL CHARACTERISTICS Ty 25 C Viy 51 Ve 1 5V Boost Viy 5V switch open unless otherwise noted PARAMETER CONDITIONS MIN TYP MAX UNITS Switch On Resistance Note 7 law 4 5A 0 07 0 1 Q e 0 13 Q Maximum Switch Duty Cycle Vrg 2 1V or Vsense 2 9V 90 93 e 86 93 Switch Frequency Vc Set to Give 50 Duty Cycle 460 500 540 kHz e 440 560 kHz Switch Frequency Line Regulation 4 3V lt Vy lt 15V e 0 0 15 Frequency Shifting Threshold on FB Pin Af 10kHz 9 0 8 1 0 1 3 V Minimum Input Voltage Note 3 e 4 0 4 3 V Minimum Boost Voltage Note 4 law lt 4 5A e 2 3 3 0 V Boost Current Note 5 Igy 1 20 35 low 4 5 90 140 mA Input Supply Current Note 6 e 3 8 5 4 mA Shutdown Supply Current VsHon OV Vsw OV Vc Open 15 50 uA e 75 pA Lockout Threshold Vc Open e 2 3 2 38 246 V Shutdown Thresholds Vc Open Device Shutting Down 9 0 13 0 37 0 60 V Device Starting Up e 0 25 0 45 0 7 V Sync
17. UT MAX is equal to s 908 8 Noo ad Note that there is less load current available at the higher input voltage because inductor ripple current increases This is not always the case Certain combinations of inductor value and input voltage range may yield lower available load current at the lowest input voltage due to reduced peak switch current at high duty cycles If load current is close to the maximum available please check maximum available current at both input voltage ex tremes To calculate actual peak switch current with a given set of conditions use Vour Vin Vour Isw PEAK lour LT 1506 APPLICATIONS INFORMATION CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR For most applications the output inductor will fall in the range of 3uH to 20H Lower values are chosen to reduce physical size of the inductor Higher values allow more output current because they reduce peak current seen by the LT1506 switch which has a 4 5A limit Higher values also reduce output ripple voltage and reduce core loss Graphs inthe Typical Performance Characteristics section show maximum output load current versus inductor size and input voltage A second graph shows core loss versus inductor size for various core materials When choosing an inductor you might have to consider maximum load current core and copper losses allowable component height output voltage ripple EMI fault cur rent in the inductor saturation
18. an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation Application Note 19 has more details on the theory of slope compensation At power up when Vc is being clamped by the FB pin see Figure 2 Q2 the sync function is disabled This allows the frequency foldback to operate in the shorted output con dition During normal operation switching frequency is controlled by the internal oscillator until the FB pin reaches 1 5V after which the SYNC pin becomes operational THERMAL CALCULATIONS Power dissipation in the LT1506 chip comes from four sources switch DC loss switch AC loss boost circuit current and input quiescent current The following 17 LT 1506 APPLICATIONS INFORMATION formulas show how to calculate each of these losses These formulas assume continuous mode operation so they should not be used for calculating efficiency at light load currents Switch loss E Rew lour vour IN Boost current loss Vour lour 50 Vin Quiescent current loss 24ns Iour Vin 0 001 Voyr 0 005 7 IN Rew Switch resistance 0 07 24ns Equivalent switch current voltage overlap time f Switch frequency Example with 10V Voyt 5V and lour Pew ono 24 107 10 500 10 5 8 50 2 Po toon sans EHC 0 04W Total power dissipation is 0 68 0 15 0 04 0
19. and of course cost The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements 1 Choose a value in microhenries from the graphs of maximum load currentand core loss Choosing a small inductor with lighter loads may result in discontinuous mode of operation butthe LT1506 is designed to work well in either mode Keep in mind that lower core loss means higher cost at least for closed core geometries like toroids The core loss graphs show absolute loss for a 3 3V output so actual percent losses must be calculated for each situation Assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions If maxi mum load current is 0 5A for instance a 0 5A inductor may not survive a continuous 4 5A overload condition Dead shorts will actually be more gentle on the induc tor because the LT1506 has foldback current limiting 2 Calculate peak inductor current at full load current to ensure that the inductor will not saturate Peak current can be significantly higher than output current espe cially with smaller inductors and lighter loads so don t omit this step Powdered iron cores are forgiving because they saturate softly whereas ferrite cores saturate abruptly Other core materials fall in between somewhere The following formula assumes continu ous mode of operation but it errs only slightly on th
20. ange of inductor types and can tell you about the latest devel opments in low profile surface mounting etc 10 LT 1506 APPLICATIONS INFORMATION Table 2 SERIES CORE VENDOR DC RESIS MATER HEIGHT PART NO uH Amps TYPE TANCE Q IAL mm Coiltronics CTX2 1 2 4 1 Tor 0 011 KMu 4 2 CTX5 4 5 44 Tor 0 019 KMu 6 4 CTX8 4 8 3 5 Tor 0 020 KMu 6 4 CTX2 1P 2 3 4 Tor 0 014 52 4 2 CTX2 3P 2 4 6 Tor 0 012 52 48 CTX5 4P 5 3 3 Tor 0 027 52 6 4 Sumida CDRH125 10 4 0 SC 0 025 Fer CDRH125 12 3 5 SC 0 027 Fer CDRH125 15 3 3 SC 0 030 Fer CDRH125 18 3 0 SC 0 034 Fer Coilcraft DT3316 222 22 5 SC 0 035 Fer 5 1 DT3316 332 3 3 5 SC 0 040 Fer 5 1 DT3316 472 4 7 3 SC 0 045 Fer 9 1 Pulse PE 53650 4 4 8 Tor 0 017 Fer 9 1 PE 53651 5 54 Tor 0 018 Fer 9 1 PE 53652 9 5 5 Tor 0 022 Fer 10 PE 53653 16 9 1 Tor 0 032 Fer 10 Dale IHSM 4825 2 7 5 1 Open 0 034 Fer 5 6 IHSM 4825 47 4 0 Open 0 047 Fer 5 6 IHSM 5832 10 4 3 Open 0 053 Fer 7 1 IHSM 5832 15 3 5 Open 0 078 Fer 7 1 IHSM 7832 22 3 8 Open 0 054 Fer 7 1 Tor Toroid SC Semiclosed geometry Fer Ferrite core material 52 Type 52 powdered iron core material KMu Kool Mu Output Capacitor The output capacitor is normally chosen by its Effective Series Resistance ESR because this is what
21. are not normally a problem but at very low input voltage they may cause erratic operation because the input voltage drops below the minimum specification Problems can also occur if the input to output voltage differential is near minimum The amplitude of these dips is normally a function of capacitor ESR and ESL because the capacitive reactance is small compared to these terms ESR tends to be the dominate term and is inversely related to physical capacitor size within a given capacitor type SYNCHRONIZING SYNC Option for DD Package The SYNC pin is used to synchronize the internal oscilla tor to an external signal The SYNC input must pass from a logic level low through the maximum synchronization threshold with a duty cycle between 10 and 90 The input can be driven directly from a logic level output The synchronizing range is equal to initial operating frequency up to 1MHz This means that minimum practical sync frequency is equal to the worst case high self oscillating frequency 560kHz not the typical operating frequency of 500kHz Caution should be used when synchronizing above 700kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced This type of subharmonic switching only occurs at input voltages less than twice output voltage Higher inductor values will tend to eliminate this problem See Frequency Compensation section for a discussion of
22. d Standard surface mount external parts are used including the inductor and capacitors There is the optional function of shutdown or synchronization A shutdown signal reduces supply current to 20 Synchronization allows an external logic level sig naltoincreasetheinternal oscillator from 580kHzto 1 MHz 47 and LT are registered trademarks of Linear Technology Corporation TYPICAL APPLICATION Efficiency vs Load Current 5V to 3 3V Down Converter OUTPUT 3 3V EFFICIENCY S OPEN LT1506 3 3 SHDN SENSE GND Vc 50uF T OR CERAMIC Man C1 100uF 10V SOLID 70 TANTALUM 0 05 10 15 20 25 30 35 40 LOAD CURRENT A 1506 01 1506 TA02 LT 1506 ABSOLUTE MAXIMUM RATINGS put WONBOG uisi oro RE RIEN IDU BUE trei m 16V 7V BOOST Pin Above Input Voltage 15V Operating Junction Temperature Range SHDN Pi VOlaggnsniuesieiteci toc ipse 7V ETTSDBU 0 to 125 C FB Pin Voltage Adjustable Part 3 5V ETIBSDBI uei sterii 40 to 125 C FB Pin Current Adjustable Part mA Storage Temperature Range 65 C to 150 C Sense Voltage Fixed 3 3V
23. e high side for discontinuous mode so it can be used for all conditions Mor our PEAK 7 lour ZUM Vin Maximum input voltage f Switching frequency 500kHz Decideifthe design can tolerate an open core geom etry like a rod or barrel which have high magnetic field radiation or whether it needs a closed core like a toroid to prevent EMI problems One would not want an open core next to a magnetic storage media for instance This isatough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radia tion will be a problem Start shopping for an inductor see representative surface mount units in Table 2 which meets the requirements of core shape peak current to avoid saturation average current to limit heating and fault current if the inductor gets too hot wire insulation will melt and cause turn to turn shorts Keep in mind that all good things like high efficiency low profile and high temperature operation will increase cost sometimes dramatically Get a quote on the cheapest unit first to calibrate yourself on price then ask for what you really want After making an initial choice consider the secondary things like output voltage ripple second sourcing etc Use the experts in the Linear Technology s applica tions department if you feel uncertain about the final choice They have experience with a wide r
24. ess the ratio of inputto output voltage exceeds 3 4 1 The only reason to consider a larger diode is the worst case condition of a high input voltage and overloaded not shorted output Under short circuit conditions foldback current limit will reduce diode current to less than 2 6A but if the output is overloaded 12 LT 1506 APPLICATIONS INFORMATION and does not fall to less than 1 3 of nominal output voltage foldback will not take effect With the overloaded condi tion output current will increase to a typical value of 5 7A determined by peak switch current limit of 6A With Vin 15V Voyt 4V 5V overloaded and Igyt 5 7A 5 7 15 4 Ip ave 4 18 This is safe for short periods of time but it would be prudent to check with the diode manufacturer if continu Ous operation under these conditions must be tolerated BOOST PIN CONSIDERATIONS For most applications the boost components are a 0 27uF capacitor and a 1N914 or 1N4148 diode The anode is connected to the regulated output voltage and this gener ates a voltage across the boost capacitor nearly identical to the regulated output In certain applications the anode may instead be connected to the unregulated input volt age This could be necessary if the regulated output voltage is very low lt 3V or if the input voltage is less than 5V Efficiency is not affected by the capacitor value but the capacitor should have an ESR of less than 1Q to ensure
25. f capacitor used at the input to regulators Aluminum electrolytics are lowest cost but are physically large to achieve adequate ripple current rating and size con Straints especially height may preclude their use Ceramic capacitors are now available in larger values and their high ripple current and voltage rating make them ideal for input bypassing Cost is fairly high and footprint may also be somewhat large Solid tantalum capacitors would be a good choice except that they have a history of occasional spectacular failures when they are subjected to large current surges during power up The capacitors can short and then burn with a brilliant white light and lots of nasty smoke This phenomenon occurs in only a small percentage of units but it has led some OEM companies to forbid their use in high surge applications The input bypass capacitor of regulators can see these high surges when a battery or high capacitance source is connected Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability AVX TPS series for instance see Table 3 but even these units may fail if the input voltage surge approaches the maximum voltage rating of the capacitor AVX recom mends derating capacitor voltage by 2 1 for high surge applications Larger capacitors may be necessary when the input volt age is very close to the minimum specified on the data sheet Small voltage dips during switch on time
26. hronization Threshold e 1 5 2 2 V Synchronizing Range 580 1000 kHz SYNC Pin Input Resistance 40 kQ The denotes specifications which apply over the full operating temperature range Note 1 Absolute Maximum Ratings are those values beyond which the life of a device may be impaired Note 2 Gain is measured with a Vc swing equal to 200mV above the switching threshold level to 200mV below the upper clamp level Note 3 Minimum input voltage is not measured directly but is guaranteed by other tests It is defined as the voltage where internal bias lines are still regulated so that the reference voltage and oscillator frequency remain constant Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current See Applications Information Note 4 This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch Note 5 Boost current is the current flowing into the boost pin with the pin held 5V above input voltage It flows only during switch on time Note 6 Input supply current is the bias current drawn by the input pin with switching disabled Note 7 Switch on resistance is calculated by dividing Vix to Ve voltage by the forced current 4 5A See Typical Performance Characteristics for the graph of switch voltage at other currents Note 8 Transconductance and voltage gain refer to the internal amplifier exclusive of the voltage
27. impedance path to maintain an equal voltage swing in L1B improving regulation In a flybackconverter during switch ontime allthe converter s energy is stroed in L1A only since no current flows in L 1B At switch off energy is transferred by magnetic coupling into L1B powering the 5V rail C4 pulls L1B positive during switch on time causing currentto flow and energy to build in L1B and C4 At switch off the energy stored in both L1B and C4 supply the 5V rail This reduces the current in L1A and changes L1B current waveform from square to triangular For details on this circuit see Design Note 100 INPUT OUTPUT 6V TO 15V 5V LT1506 SHDN GND Vc ci 100uF 10V TANT C3 10uF 25V CERAMIC GND m C5 C4 Li 100uF 10V TANT L1 IS A SINGLE CORE WITH TWO WINDINGS OUTPUT BH ELECTRONICS 501 0726 5V TOKIN IE475ZY5U C304 D3 iso ds t IF LOAD CAN GO TO ZERO AN OPTIONAL PRELOAD OF 1k TO 5k MAY BE USED TO IMPROVE LOAD REGULATION D1 D3 MBRD340 Figure 17 Dual Output SEPIC Converter AL M Information furnished by Linear Technology Corporation is believed to be accurate and reliable However no responsibility is assumed for its use Linear Technology Corporation makes no represen tation thatthe interconnection of its circuits as described herein will not infringe on existing patent rights 23 LT 1506 PACKAGE DESCRIPTION R Package Dimensions in inches millimeters unless other
28. in parallel with the Rc Cc network on the Vc pin Pole frequency for this capacitor is typically set at one fifth of switching frequency so that it provides significant attenuation of switching ripple but does not add unacceptable phase shift at loop unity gain frequency With Rc 3k 9 9 2n Ro 2x 500 10 531 How Do Test Loop Stability The standard compensation for LT1506 is a 1 5nF capacitor for Cc with Rc 0 While this compensation will work for most applications the optimum value for loop compensation components depends to various extent on parameters which are not well controlled These include inductor value 30 due to production tolerance load current and ripple current variations output capacitance 20 to 50 due to production tolerance tempera ture aging and changes at the load output capacitor ESR 200 due to production tolerance temperature and aging and finally DC input voltage and output load current This makes it important for the designer to check out the final design to ensure that itis robust and tolerant of all these variations check switching regulator loop stability by pulse loading the regulator output while observing transient response at the output using the circuit shown in Figure 13 The regulator loop is hit with a small transient AC load current at a relatively low frequency 50Hz to 1kHz This causes the output to jump afew millivol
29. ixed voltage 3 3 parts have the divider included on the chip and the FB pin is used as a SENSE pin connected directly to the 3 3V output Three additional functions are performed by the FB pin When the pin voltage drops below 1 7V switch current limit is reduced Below 1 5V the external sync function is disabled Below 1V switching frequency is also reduced See Feedback Pin Function section in Applica tions Information for details BOOST The BOOST pin is used to provide a drive voltage higher than the input voltage to the internal bipolar NPN power switch Without this added voltage the typical switch voltage loss would be about 1 5V The additional boost voltage allows the switch to saturate and voltage loss approximates that of a 0 07 FET structure but with much smaller die area Efficiency improves from 75 for conventional bipolar designs to gt 89 for these new parts Vin This is the collector of the on chip power NPN switch This pin powers the internal circuitry and internal regula tor At NPN switch on and off high dl dt edges occur on this pin Keep the external bypass and catch diode close to this pin All trace inductance on this path will create a voltage spike at switch off adding to the Vcr voltage across the internal NPN GND The GND pin connection needs consideration for two reasons First itacts as the reference for the regulated output so load regulation will suffer if the ground end of the l
30. l case subharmonic switching occurs as evidenced by alternat ing pulse widths seen at the switch node In more severe cases the regulator squeals or hisses audibly even though the output voltage is still roughly correct None of this will show on a theoretical Bode plot because Bode is an amplitude insensitive analysis Tests have shown that if ripple voltage on the Vc is held to less than 100mVp p the LT1506 will be well behaved The formula below will give an estimate of Vc ripple voltage when is added to the loop assuming that Rc is large compared to the reactance of Cc at 500kHz Rc Gus Vour ESRJ 2 4 wot Gma Error amplifier transconductance 2000uMho Vo RIPPLE If a computer simulation of the LT1506 showed that a series compensation resistor of 3k gave best overall loop response with adequate gain margin the resulting Vc pin ripple voltage with Viy 10V Vout 5V ESR 0 10 L 10uH would be 3k 2 107 10 5 o 1 24 oJ to 1075 500 10 This ripple voltage is high enough to possibly create subharmonic switching In most situations a compromise value 2k in this case for the resistor gives acceptable phase margin and no subharmonic problems In other 0 144V VefRiPPLE cases the resistor may have to be larger to get acceptable phase response and some means must be used to control ripple voltage at the Vc pin The suggested way to do this isto add a capacitor Cp
31. lop is reset and the switch turns off Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point This technique means that the error amplifier commands current to be delivered to the output rather than voltage A voltage fed system will have low phase shift up to the resonant frequency of the inductor 2 9V BIAS REGULATOR 0 012 INPUT _ INTERNAL Voc SLOPE COMP 500kHz sync OSCILLATOR SHUTDOWN COMPARATOR SHDN CO a CURRENT SENSE AMPLIFIER VOLTAGE GAIN 20 CURRENT COMPARATOR FOLDBACK CURRENT LIMIT CLAMP and output capacitor then an abrupt 180 shift will occur The current fed system will have 90 phase shift ata much lower frequency but will not have the additional 90 shift until well beyond the LC resonant frequency This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response High switch efficiency is attained by using the BOOST pin to provide a voltage to the switch driver which is higher than the input voltage allowing switch to be saturated This boosted voltage is generated with an external capaci tor and diode Two comparators are connected to the shutdown pin One has a2 38V threshold for undervoltage lockout and the second has a 0 4V threshold for complete shutdown BOOST Q1 POWER SWITCH FREQUENCY PARASITIC DIODES S
32. nd 15V Vy the theoretical summed output ripple current cancels completely To reduce board space and ripple voltage C1 and C3 are ceramic capacitors Loop compensation C4 must be adjusted when using ceramic output capacitors due to the lack of effective series resis tance The typical tantalum compensation of 1 5nF is increased to 22nF x3 for the ceramic output capacitor If synchronization is not used and the internal oscillators free run the circuit will operate correctly but ripple cancellation will not occur Input and output capacitors must be ripple rated for the total output current C1 MARCON THCS50E1E106Z D1 ROHM RB051L 40 LT1506 SYNC Vc SYNC SW GND Vy BOOST FB 5V 12A C1 10uF 25V Figure 15 Current Sharing 12A Supply 22 LT 1506 APPLICATIONS INFORMATION Redundant Operation The circuit shown in Figure 15 is fault tolerant when Operating at less than 8A of output current If one device fails the output will remain in regulation The feedback loop will compensate by raising the voltage on the Vc pin increasing switch current of the two remaining devices BUCK CONVERTER WITH ADJUSTABLE SOFT START Large capacitive loads can cause high input currents at start up Figure 16 shows a circuit that limits the dv dt of the output at start up controlling the capacitor charge rate The buck converter is a typical configuration with the addition of R3 R4 Css and Q1 As the output starts to rise Q1
33. nt ripple which can have an RMS value as high as one half of load current Ripple current ratings on the capacitor must be observed to ensure reliable operation In many cases itis necessary to parallel two capacitors to obtain the required ripple rating Both capacitors must be of the same value and manufacturer to guarantee power sharing The actual value of the capacitor in microfarads is not particularly important because at 500kHz any value above SuF is essentially resistive RMS ripple current rating is the critical parameter Actual RMS current can be calculated from 2 IRIPPLE RMS bur 16 LT 1506 APPLICATIONS INFORMATION The term inside the radical has a maximum value of 0 5 when input voltage is twice output and stays near 0 5 for a relatively wide range of input voltages It is common practice therefore to simply use the worst case value and assume that RMS ripple currentis one half of load current At maximum output current of 4 5A for the LT1506 the input bypass capacitor should be rated at 2 25A ripple current Note however that there are many secondary considerations in choosing the final ripple current rating These include ambient temperature average versus peak load current equipment operating schedule and required product lifetime For more details see Application Notes 19 and 46 and Design Note 95 Input Capacitor Type Some caution must be used when selecting the type o
34. nterplane or surface capaci tance to the switching nodes are minimized If high resistor values are used the shutdown pin should be bypassed with a 1000pF capacitor to prevent coupling problems from the switch node If hysteresis is desired in the undervoltage lockout point a resistor Reg can be added to the output node Resistor values can be calcu lated from Ro Vs 2 38 AV Ngyr 1 AV 2 38 R2 3 5uA Rrg AV 25k suggested for Rio Vi Input voltage at which switching stops as input voltage descends to trip level AV Hysteresis in input voltage level Example output voltage is 5V switching is to stop if input voltage drops below 6V and should not restart unless input rises back to 7 5V AV is therefore 1 5V and Viy OV Let Rig 25k 25 6 2 38 1 5 5 1 1 5 2 38 25k 3 5pA 25k 5 2 zag Rrg 48 5 1 5 160 SWITCH NODE CONSIDERATIONS For maximum efficiency switch rise and fall times are made as short as possible To prevent radiation and high frequency resonance problems proper layout of the com ponents connected to the switch node is essential B field magnetic radiation is minimized by keeping catch diode switch pin and input bypass capacitor leads as short as possible E field radiation is kept low by minimizing the length and area of all traces connected to the switch pin and BOOST pin A ground plane should always be used under the switcher circuitr
35. oad is not at the same voltage as the GND pin of the IC This condition will occur when load current or other currents flow through metal paths between the GND pin and the load ground point Keep the ground path short between the GND pin and the load and use a ground plane when possible The second consideration is EMI caused by GND pin current spikes Internal capacitance between the Vsw pin and the GND pin creates very narrow 10ns current spikes in the GND pin If the GND pin is connected to system ground with a long metal trace this trace may radiate excess EMI Keep the path between the input bypass and the GND pin short The GND pin of the SO 8 package is directly attached to the internal tab This pin should be attached to a large copper area to improve thermal resistance Vsw The switch pin is the emitter of the on chip power NPN switch This pin is driven up to the input pin voltage during switch on time Inductor current drives the switch pin negative during switch off time Negative voltage is clamped with the external catch diode Maximum negative switch voltage allowed is 0 8V SYNC The sync pin is used to synchronize the internal oscillator to an external signal It is directly logic compat ible and can be driven with any signal between 10 and 90 duty cycle The synchronizing range is equal to initial operating frequency up to 1 MHz This pin replaces SHDN on SYNC option parts See Synchronizing section in Applicati
36. ol is a registered trademark of Magnetics Inc Metglas is a registered trademark of AlliedSignal Inc FREQUENCY kHz Switching Frequency 550 540 530 520 510 500 LI Ee 490 480 470 460 450 50 25 0 25 50 75 TEMPERATURE C 1506 G11 Maximum Load Current at Vout 3 3V 44 L 10uH 4 2 LOAD CURRENT A THRESHOLD VOLTAGE V 4 0 3 8 3 6 3 4 3 2 4 6 8 10 12 14 INPUT VOLTAGE V 1506 G13 Vc Pin Shutdown Threshold SHUTDOWN 25 0 25 50 75 100 JUNCTION TEMPERATURE C 125 1506 G15 See More Than Just Voltage Feedback in the Applications Information section 1 0 k Inductor Core Loss for 3 3V Output BOOST PIN CURRENT mA SWITCH VOLTAGE mV 500 450 400 TYPE 52 Kool Mu PERMALLOY ze w 125 2 4 6 8 10 INDUCTANCE uH 1506 G01 BOOST Pin Current DUTY CYCLE 100 0 1 2 3 4 5 SWITCH CURRENT A 1506 G14 Switch Voltage Drop 1 2 3 4 5 SWITCH CURRENT A 1506 G16 LT 1506 PIN FUNCTIONS FB SENSE The feedback pin is used to set output voltage using an external voltage divider that generates 2 42V at the pin with the desired output voltage The f
37. ons Information for details When not in use this pin should be grounded SHDN The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes Actually this pin has two separate thresholds one at 2 38V to disable switching and a second at 0 4V to force complete micropower shutdown The 2 38V threshold functions as an accurate undervoltage lockout UVLO This is sometimes used to prevent the regulator from operating until the input votlage has reached a predeter mined level Vc The Vc pin is the output of the error amplifier and the input of the peak switch current comparator It is normally used for frequency compensation but can do double duty as a current clamp or control loop override This pin sits at about 1V for very light loads and 2V at maximum load It can be driven to ground to shut off the regulator but if driven high current must be limited to 4mA LT 1506 BLOCK DIAGRAM The LT1506 is a constant frequency current mode buck converter This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch In addition to the normal error amplifier there is a current sense amplifier that monitors switch current on a cycle by cycle basis A switch cycle starts with an oscilla tor pulse which sets the Rs flip flop to turn the switch on When switch current reaches a level set by the inverting input of the comparator the flip f
38. orking correctly start varying load current and input voltage to see if can find any combination that makes the transient response look suspiciously ringy This procedure may lead to an adjustment for best loop stability or faster loop transient response Nearly always you will find that loop response looks better if you add in several for Rc Do this only if necessary because as explained before Rc above 1k may require the addition of Cr to control Vc pin ripple If everything looks OK use a heat gun and cold spray on the circuit especially the output capacitor to bring out any temperature dependent characteristics Keep in mind that this procedure does not take initial component tolerance into account You should see fairly clean response under all load and line conditions to ensure that component variations will not cause problems One note here according to Murphy the component most likely to be changed in production is the output capacitor because that is the component most likely to have manu facturer variations in ESR large enough to cause prob lems It would be a wise move to lock down the sources of the output capacitor in production A possible exception to the clean response rule is at very light loads as evidenced in Figure 14 with lj 50mA Switching regulators tend to have dramatic shifts in loop response at very light loads mostly because the inductor current becomes discontinuous One common
39. result is very slow but stable characteristics A second possibility is low phase margin as evidenced by ringing at the output with transients The good news is that the low phase margin at light loads is not particularly sensitive to component varia tion so if it looks reasonable under a transient test it will probably not be a problem in production Note that fre quency of the light load ringing may vary with component tolerance but phase margin generally hangs in there CURRENT SHARING MULTIPHASE SUPPLY The circuit in Figure 15 uses multiple LT1506s to produce a 5V 12A power supply There are several advantages to using a multiple switcher approach compared to a single larger switcher The inductor size is considerably reduced Three 4A inductors store less energy 12 2 than one 12 coil so are far smaller In addition synchronizing three 21 LT 1506 APPLICATIONS INFORMATION converters 120 out of phase with each other reduces input and output ripple currents This reduces the ripple rating size and cost of filter capacitors Current Sharing Split Input Supplies Current sharing is accomplished by joining the Vc pins to a common compensation capacitor The output of the error amplifier is stage so any number of devices be connected together The effective gm of the composite error amplifier is the multiple of the individual devices In Figure 15 the compensation capacitor C4 has been
40. ristics This is done to control power dissipation in both the IC and in the external diode and inductor during short circuit conditions A shorted output requires the switching regu lator to operate at very low duty cycles and the average current through the diode and inductor is equal to the short circuit current limit of the switch typically 6A for the LT1506 folding backto less than 3A Minimum switch on time limitations would prevent the switcher from attaining a sufficiently low duty cycle if switching frequency were maintained at 50OkHz so frequency is reduced by about 5 1 when the feedback pin voltage drops below 1V see Frequency Foldback graph This does not affect operation with normal load conditions one simply sees a gear shift in switching frequency during start up as the output voltage rises In addition to lower switching frequency the LT1506 also operates at lower switch current limit when the feedback pin voltage drops below 1 7V Q2 in Figure 2 performs this function by clamping the Vc pin to a voltage less than its normal 2 1V upper clamp level This foldback current limit greatly reduces power dissipation in the IC diode and inductor during short circuit conditions External synchro nization is also disabled to prevent interference with foldback operation Again it is nearly transparent to the user under normal load conditions The only loads that may be affected are current source loads which maintain full load c
41. rough the inductor into the output capacitor is Vour in Vour 0 For high frequency switchers the sum of ripple current slew rates may also be relevant and can be calculated from Peak to peak output ripple voltage is the sum of a triwave created by peak to peak ripple current times ESR and a square wave created by parasitic inductance ESL and ripple current slew rate Capacitive reactance is assumed to be small compared to ESR or ESL lp ESR sz Example with Viy z10V Vout 5V L 10uH ESR 2 0 10 ESL 10nH 510 5 0 5 1o 10 1075 00 10 z 10 _ ipe dt 10 1078 0 5a 0 1 10 107 109 0 05 0 01 60mVp p AT lout 1A 20 20mV DIV Vout AT lout 50mA INDUCTOR CURRENT AT lour 1 0 5A DIV INDUCTOR CURRENT AT lgyr 50mA 0 5us DIV 1374 F03 Figure 3 LT1506 Ripple Voltage Waveform CATCH DIODE The suggested catch diode D1 is a 1N5821 Schottky or its Motorola equivalent MBR330 It is rated at average forward current and 30V reverse voltage Typical forward voltage is 0 5V at 3A The diode conducts current only during switch off time Peak reverse voltage is equal to regulator input voltage Average forward current in normal operation can be calculated from lour Vin Your Vin Ip ava This formula will not yield values higher than 3A with maximum load current of 4 25A unl
42. t for a buck converter is limited by the maximum switch current rating Ip of the LT1506 This current rating is 4 5A up to 50 duty cycle DC decreasing to 3 7A at 80 duty cycle This is shown graphically in Typical Performance Characteristics and as shown in the formula below Ip 4 5 for DC lt 50 Ip 3 21 5 95 DC 6 75 DC for 50 lt DC lt 90 DC Duty cycle Voyz Vin Example with Vout 5V Vin 8V DC 5 8 0 625 and Isw max 3 21 5 95 0 625 6 75 0 625 4 3A Current rating decreases with duty cycle because the LT1506 has internal slope compensation to prevent cur rent mode subharmonic switching For more details read Application Note 19 The LT1506 is a little unusual in this regard because it has nonlinear slope compensation which gives better compensation with less reduction in current limit Maximum load current would be equal to maximum switch current for an infinitely large inductor but with finite inductor size maximum load current is reduced by one half peak to peak inductor current The following formula assumes continuous mode operation implying that the term on the right is less than one half of Ip Vous JV Uf Wu For the conditions above and L 3 3uH 5 8 5 2 3 35107 500 10 8 4 3 0 57 3 73 loUT MAX Continuous Mode 4 3 At Viy 15V duty cycle is 33 so Ip is just equal to a fixed 4 5A and lo
43. that it can be recharged fully under the worst case condi tion of minimum input voltage Almost any type of film or ceramic capacitor will work fine For nearly all applications a 0 27 pF boost capacitor works just fine but for the curious more details are provided here The size of the boost capacitor is determined by switch drive current requirements During switch on time drain current on the capacitor is approximately Igy7 50 At peak load current of 4 25A this gives atotal drain of 85mA Capacitor ripple voltage is equal to the product of on time and drain current divided by capacitor value AV toy 85mA C To keep capacitor ripple voltage to less than 0 6V a slightly arbitrary number at the worst case condition of toy 1 8us the capacitor needs to be 0 27uF Boost capacitor ripple voltage is not a critical parameter but if the minimum voltage across the capaci tor drops to less than 3V the power switch may not saturate fully and efficiency will drop An approximate formula for absolute minimum capacitor value is lour 50 Vour Viu Vour 2 8V f Switching frequency Vout Regulated output voltage Vin Minimum input voltage MIN This formula can yield capacitor values substantially less than 0 27uF but it should be used with caution since it does not take into account secondary factors such as capacitor series resistance capacitance shift with tem perature and output overload SHUTDOWN FUNCTION
44. tors fail during very high turn on surges which do not occur at the output of regulators High discharge surges such as when the regulator output is dead shorted do not harm the capacitors Unlike the input capacitor RMS ripple current in the Output capacitor is normally low enough that ripple cur rent rating is not an issue The current waveform is triangular with a typical value of 200mAgys The formula to calculate this is Output Capacitor Ripple Current RMS 0 29 Vour Mn IniPPLE RMS 11 LT 1506 APPLICATIONS INFORMATION Ceramic Capacitors Higher value lower cost ceramic capacitors are now available in smaller case sizes These are ideal for input bypassing because of their high ripple rating and tolerance to turn on surges As output capacitors caution must be used Solid tantalum capacitor s ESR generates a loop Zero at 5kHz to 50kHz that is beneficial in giving accept able loop phase margin Ceramic capacitors remain ca pacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective When using ceramic output capacitors the loop compensation pole frequency must be reduced by a typical factor of 10 OUTPUT RIPPLE VOLTAGE Figure 3 shows a typical output ripple voltage waveform for the LT1506 Ripple voltage is determined by the high frequency impedance of the output capacitor and ripple current through the inductor Peak to peak ripple current th
45. ts then settle back to the original value as shown in Figure 14 A well behaved loop will settle back cleanly whereas a loop with poor phase or gain margin will ring as it settles The number of rings indicates the degree of stability and the frequency of the ringing shows the approximate unity gain fre quency of the loop Amplitude of the signal is not particu larly important as long as the amplitude is not so high that the loop behaves nonlinearly 20 LT 1506 APPLICATIONS INFORMATION SWITCHING REGULATOR ADJUSTABLE S ADJUSTABLE INPUT SUPPLY 100uF TO 1000uF TO 100Hz TO 1kHz 100mV TO 1Vp p OSCILLOSCOPE SYNC RIPPLE FILTER TO X1 OSCILLOSCOPE PROBE 1506 F13 Figure 13 Loop Stability Test Circuit Vour AT 500mA BEFORE FILTER OUT Vout AT 500mA AFTER B Vou ATI AFTER FI OUT 10mV TER our 50mA LTER LOAD PULSE THROUGH 50Q f 780Hz 0 2ms DIV 1375 76 F14 Figure 14 Loop Stability Check The output of the regulator contains both the desired low frequency transient information and a reasonable amount of high frequency 500kHz ripple The ripple makes it difficult to observe the small transient so a two pole 100kHz filter has been added This filter is not particularly critical even if it attenuated the transient signal slightly this wouldn t matter because amplitude is not critical After verifying that the setup is w
46. urrent with output voltage less than 50 offinal value In these rare situations the feedback pin can be clamped above 1 5V with an external diode to defeat foldback cur rent limit Caution clamping the feedback pin means that frequency shifting will also be defeated so a combination of high input voltage and dead shorted output may cause the LT1506 to lose control of current limit LT 1506 APPLICATIONS INFORMATION The internal circuitry which forces reduced switching frequency also causes current to flow out of the feedback pin when output voltage is low The equivalent circuitry is shown in Figure 2 Q1 is completely off during normal operation If the FB pin falls below 1V Q1 begins to conduct current and reduces frequency at the rate of approximately 5kHz pA To ensure adequate frequency foldback under worst case short circuit conditions the external divider Thevinin resistance must be low enough to pull 150pA out of the FB pin with 0 6V on the pin lt 4k The net result is that reductions in frequency and current limit are affected by output voltage divider imped ance Although divider impedance is not critical caution should be used if resistors are increased beyond the suggested values and short circuit conditions will occur with high input voltage High frequency pickup will increase and the protection accorded by frequency and current foldback will decrease MAXIMUM OUTPUT LOAD CURRENT Maximum load curren
47. ut to ground Gain rolls off smoothly above the 600Hz pole frequency set by the 100uF output capacitor Phase drop is limited to about 70 Phase recovers and gain levels off at the zero fre quency 16kHz set by capacitor ESR 0 10 CURRENT MODE OUTPUT 1506 F09 18 LT 1506 APPLICATIONS INFORMATION 40 40 Vin 10V lt 7 Vout 5V GAI QUT 2A E gt d 20 0 lt E e o 40 d e a _ A E PHASE E z od 20 80 5 E 40 120 10 100 1k 10k 100k 1M FREQUENCY Hz 1505 F10 Figure 10 Response from Vc Pin to Output Erroramplifiertransconductance phase and gain are shown in Figure 11 The error amplifier can be modeled as a transconductance of 2000uMho with an output imped ance of 200kQ in parallel with 12pF In all practical applications the compensation network from Vg to ground has a much lower impedance than the output impedance of the amplifier at frequencies above 500Hz This means that the error amplifier characteristics them selves do notcontribute excess phase shiftto the loop and the phase gain characteristics of the error amplifier sec tion are completely controlled by the external compensa tion network In Figure 12 full loop phase gain characteristics are shown with a compensation capacitor of 1 5nF giving the error amplifier a pole at 530Hz with phase rolling
48. wise noted 7 Lead Plastic DD Pak LTC DWG 05 08 1462 0 060 1 524 0 390 0 415 0 256 20 060 9 906 10 541 0 165 0 180 i 4 191 4 572 p 15 i P F3 0 060 0183 0 330 0 370 0 059 0004 1 524 4 648 8 382 9 398 0 102820 NN 0 095 0 115 ds 2 413 2 921 m 0 040 0 060 0 050 0 012 V gt 0 143 0012 SLOT 06 188 a o _ 1 270 0 305 BOTTOM VIEW OF DD PAK 268210308 gt 0 660 0 914 HATCHED AREA IS SOLDER PLATED COPPER HEAT SINK S8 Package 8 Lead Plastic Small Outline Narrow 0 150 LTC DWG 05 08 1610 0189 0197 4 801 5 004 n P eum 0 008 0 010 0 203 0 254 g i TYP Eee FO 0016 0050 m ATIS Tee 1 1 cmm sucum Loo pase mcum ERN DIMENSION DOES NOT INCLUDE MOLD FLASH MOLD FLASH ORT SHALL NOT EXCEED 0 006 0 152mm PER SIDE DIMENSION DOES NOT INCLUDE INTERLEAD FLASH INTERLEAD 1 2 3 4 FLASH SHALL NOT EXCEED 0 010 0 254mm PER SIDE PART NUMBER DESCRIPTION COMMENTS LT1074
49. y to prevent interplane cou pling A suggested layout for the critical components is shown in Figure 5 Note that the feedback resistors and compensation components are kept as far as possible 14 LT 1506 APPLICATIONS INFORMATION from the switch node Also note that the high current ground path of the catch diode and input capacitor are kept very short and separate from the analog ground line The high speed switching current path is shown schemati cally in Figure 6 Minimum lead length in this path is essential to ensure clean switching and low EMI The path including the switch catch diode and input capacitor is MINIMIZE LT1506 C3 D1 LOOP the only one containing nanosecond rise and fall times If you follow this path on the PC layout you will see that it is irreducibly short If you move the diode or input capacitor away from the LT1506 get your resum in order The other paths contain only some combination of DC and 500kHz triwave so are much less critical CONNECT TO GROUND PLANE iud Vout TAKE OUTPUT DIRECTLY FROM END OF OUTPUT CAPACITOR A CONNECT TO PLACE FEEDTHROUGHS GROUND PLANE AROUND GND PIN FOR GOOD m THERMAL CONDUCTIVITY KELVIN SENSE KEEP FB AND Vc COMPONENTS Pa AWAY FROM HIGH FREQUENCY Vout 1506 FOS HIGH CURRENT COMPONENTS Figure 5 Suggested Layout Topside Only Shown SWITCH NODE O HIGH

LINEAR TECHNOLOGY LT1506 Manual(1)

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