NATIONAL SEMICONDUCTOR LM3406/06HV Manual
Contents
1. lL 0 5 X Ali In order to prevent inductor saturation the inductor s peak current rating must be above 1 8A A 15 uH off the shelf in ductor rated to 2 4A peak and 2 2A average with a DCR of 47 mQ will be used USING AN OUTPUT CAPACITOR This application does not require high frequency PWM dim ming allowing the use of an output capacitor to reduce the size and cost of the output inductor while still meeting the 2096 p p 300 mA target for LED ripple current To select the proper output capacitor the equation from Buck Regulators with Out put Capacitors is re arranged to yield the following Alf Zc Ip Al The dynamic resistance rp of one LED can be calculated by taking the tangent line to the Ve vs 1 curve in the LED datasheet Figure 8 shows an example rp calculation FORWARD VOLTAGE V 400 800 FORWARD CURRENT 1200 1600 2000 30020324 FIGURE 10 Calculating rp from the Ve vs lp Curve Extending the tangent line to the ends of the plot yields values for AV and Al of 0 7V and 2000 mA respectively Dynamic resistance is then rn AV Alp 0 7V 2A 0 350 The most filtering and therefore the highest output capaci tance is needed when Al is highest which occurs at Vin www national com 18 max Inductor ripple current with one LED is 516 The required impedance of C is calculated Zo 0 3 0 516 0 3 x 0 352. k LM3406 Ir 1 5A 5 L1 LED1 W3 1 One to Co five LEDs LEDn W3 Rsns 30020318 FIGURE 7 Schematic for Design Example 1 Ron and ton A moderate switching frequency of 500 kHz will balance the requirements of inductor size and overall power efficiency The LM3406 will allow some shift in switching frequency when Vo changes due to the number of LEDs in series so the cal culation for Roy is done at the mid point of three LEDs in series where Vo 11 8V Note that the actual Roy calculation is done with the high accuracy expression listed in the Ap pendix 1 Ron en fsw x 1 x 10 Ron 144 The closest 1 tolerance resistor is 143 The switching frequency and on time of the circuit should be checked for one three and five LEDs using the equations relating Roy and ton to fay As with the Roy calculation the actual foy and ton values have been calculated using the high accuracy ex pressions listed in the Appendix 1 fow SW 4x10 x Ron V ton 1 x 10 x Ron x VIN low 1 528 ns www national com 14 long Leps 1014 ns OUTPUT INDUCTOR Since an output capacitor will be used to filter some of the AC ripple current the inductor ripple current can be set higher than the LED ripple current A value of 40 p_p is typical in many buck converters With the target ripple current determined the inductance can be chosen V z INT
3. Current Limit vs V LM3406 5 10 15 20 25 30 35 40 45 INPUT VOLTAGE V INPUT VOLTAGE V 30020339 30020341 CURRENT LIMIT A MAXIMUM OUTPUT VOLTAGE V Current Limit vs Vy LM3406HV 2 40 5 10 15 20 25 30 35 40 45 50 55 60 INPUT VOLTAGE V 30020340 0 0 10 20 30 40 50 60 70 80 INPUT VOLTAGE V 30020342 Block Diagram 7V BIAS REGULATOR mm GND 300 ns MIN OFF TIMER Application Information THEORY OF OPERATION The LM3406 and LM3406HV are buck regulators with a wide input voltage range low voltage reference and a fast output enable disable function These features combine to make them ideal for use as a constant current source for LEDs with forward currents as high as 1 5A The controlled on time COT architecture uses a comparator and a one shot on timer that varies inversely with input and output voltage in stead of a fixed clock The LM3406 06HV also employs an integrator circuit that averages the output current When the converter runs in continuous conduction mode CCM the controlled on time maintains a constant switching frequency over changes in both input and output voltage These features combine to give the LM3406 06HV an accurate output cur rent fast transient response and constant switching frequen cy over a wide range of conditions CURRENT LIMIT OFF TIMER 30020303 CONTROLLED ON TIME OVERVIEW Figure 1 shows a simplified version of the feedback sy
4. 0 490 A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 440 kHz Co 1 2 X TT x 0 49 x 4 4 x 105 0 74 uF This calculation assumes that Co will be a ceramic capacitor and therefore impedance due to the equivalent series resis tance ESR and equivalent series inductance ESL of of the device is negligible The closest 10 tolerance capacitor val ue is 1 uF The capacitor used should have an X7R dielectric and should be rated to 50V The high voltage rating ensures that will not be damaged if the LED fails open circuit and a load dump occurs Several manufacturers produce ceramic capacitors with these specifications in the 1206 case size With only 4V of DC bias a 50V rated ceramic capacitor will have better than 90 of it s rated capacitance which is more than enough for this design Rens Using the expression for Rays Reng 0 2 1 5 0 1330 Sub 1Q resistors are available in both 1 and 5 tolerance 1 0 130 device is the closest value and a 0 33W 1210 size device will handle the power dissipation of 290 mW With the resistance selected the average value of LED current is re calculated to ensure that current is within the 5 toler ance requirement From the expression for average LED current l 0 2 0 13 1 54A 3 above the target current INPUT CAPACITOR Controlling input ripple current and voltage is critical in auto motive applications where s
5. 30020312 FIGURE 5 Low Power Shutdown THERMAL SHUTDOWN Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction temperature is ex ceeded The threshold for thermal shutdown is 165 C with a 11 25 C hysteresis both values typical During thermal shut down the MOSFET and driver are disabled www national com AH90OVEIN 1 90vV IN 1 LM3406 LM3406HV Design Considerations SWITCHING FREQUENCY Switching frequency is selected based on the trade offs be tween efficiency better at low frequency solution size cost smaller at high frequency and the range of output voltage that can be regulated wider at lower frequency Many appli cations place limits on switching frequency due to EMI sen sitivity The on time of the LM3406 06HV can be programmed for switching frequencies ranging from the 10 s of kHz to over 1 MHz This on time varies in proportion to both Vin and Vo in order to maintain first order control over switching frequen cy however in practice the switching frequency will shift in response to large swings in Vin or Vo The maximum switch ing frequency is limited only by the minimum on time and minimum off time requirements LED RIPPLE CURRENT Selection of the ripple current Aic through the LED array is similar to the selection of output ripple voltage in a standard voltage regulator Where the output ripple in a voltage regu lator is commonly 1 to 5 o
6. VINS COMPARATOR INTERNAL REGULATOR Vec REG Vcc Regulated Output lt lec lt 5 mA ERDE 8 Vs Vec Limit Vin 24V Voc OV a 20 m 3 www national com AH90OVEIN 1 90vV IN 1 LM3406 LM3406HV Symbol Conditions min Typ Unit Vcc uv TH Voc Under voltage Lock out Voc Increasing 5 3 Threshold Voc UV HYS Voc Under voltage Lock out Voc Decreasing Hysteresis liN oP I Operating Current Non switching CS 0 5V BE ij Shutdown Current RON OV zo so CURRENT LIMIT DIM COMPARATOR MOSFET AND DRIVER Aa Buck Switch On Resistance Igy 200 mA BOOT 6 3V RN X Q VDR uvLo BOOT Under voltage Lock out BOOT SW Increasing 2 9 V Threshold Vpn uvs BOOT Under voltage Lock out BOOT SW Decreasing mV Hysteresis THERMAL SHUTDOWN Tsp Thermal Shutdown Threshold Tsp Hys Thermal Shutdown Hysteresis THERMAL RESISTANCE A EZ 1 DE 2 Note 1 Absolute Maximum Ratings indicate limits beyond which damage to the device may occur including inoperability and degradation of device reliability and or performance Functional operation of the device and or non degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in the Operating Ratings is not implied The recommended Operating Ratings indicate conditions at which the device is functional and the device should not be operated beyond such conditions Note 2 The human body model
7. b support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation All other brand or product names may be trademarks or registered trademarks of their respective holders CopyrightO 2008 National Semiconductor Corporation For the most current product information visit us at www national com National Semiconductor National Semiconductor Europe National Semiconductor Asia National Semiconductor Japan Americas Technical Technical Support Center Pacific Technical Support Center Technical Support Center Support Center Email europe support 9 nsc com Email ap support 9 nsc com Email jpn feedback 9 nsc com Email support 9 nsc com German Tel 49 0 180 5010 771 Tel 1 800 272 9959 English Tel 44 0 870 850 4288 www national com
8. apply to the LED ripple cur rent Aip For a controlled on time converter such as LM3406 06HV the ripple current is described by the following expression V Ai Alf X ton The triangle wave inductor current ripple flows through Rens and produces a triangle wave voltage at the CS pin To pro vide good signal to noise ratio SNR the amplitude of CS pin ripple voltage Avcs should be at least 25 mVp p Aves is de scribed by the following Avcs Ale X Rens www national com 12 BUCK CONVERTERS WITH OUTPUT CAPACITORS A capacitor placed in parallel with the LED s can be used to reduce the LED current ripple while keeping the same aver age current through both the inductor and the LED array With an output capacitor the output inductance can be lowered making the magnetics smaller and less expensive Alterna tively the circuit could be run at lower frequency but keep the same inductor value improving the power efficiency Both the peak current limit and the OVP OCP comparator still monitor peak inductor current placing a limit on how large Ai can be even if Aic is made very small Adding a capacitor that re duces Ai to well below the target provides headroom for changes in inductance or V that might otherwise push the peak LED ripple current too high Figure 6 shows the equivalent impedances presented to the inductor current ripple when an output capacitor Co and its equivalent series resistance ESR ar
9. are capable of slewing the LED current from O to the target level fast enough For such applications the LED current slew rate can by increased by shorting the LED current with a N MOSFET placed in parallel to the LED or LED array as shown in Figure 3 While the parallel FET is on the output current flows through it effectively reducing the output voltage to equal the CS pin voltage of 0 2V This dim ming method maintains a continuous current through the inductor and therefore eliminates the biggest delay in turning the LED s or and off The trade off with parallel FET dimming is that more power is wasted while the FET is on although in most cases the power wasted is small compared to the power dissipated in the LEDs Parallel FET circuits should use no output capacitance or a bare minimum for noise filtering in order to minimize the slew rate of output voltage Dimming FET Q1 can be driven from a ground referenced source be cause the source stays at 0 2V along with the CS pin www national com AH90OVEIN 1 90v IN 1 LM3406 LM3406HV VIN VINS BOOT RON LM3406 06HV Cc UL VOUT Q1 RSNS 30020327 FIGURE 3 Dimming with a Parallel FET PEAK CURRENT LIMIT The current limit comparator of the LM3406 06HV will engage whenever the power MOSFET current equal to the inductor current while the MOSFET is on exceeds 2 14 typical The power MOSFET is disabled for a cool down time that of ap proximately 1
10. device will handle the power dissipation of 290 mW With the resistance selected the average value of LED current is re calculated to ensure that current is within the 5 toler ance requirement From the expression for average LED current 0 2 0 13 1 54A 3 above the target current INPUT CAPACITOR Following the calculations from the Input Capacitor section Av n max Will be 24V x 276p 5 480 mV The minimum re quired capacitance is calculated for the largest toj corre sponding to five LEDs As with the output capacitor this required value is low enough to use a ceramic capacitor and again the effective capaci tance will be lower than the rated value with 24V across Cy Reviewing plots of C vs DC Bias for several capacitors re veals that a 4 7 pF 1812 size capacitor in X7R rated to 50V loses about 40 of its rated capacitance at 24V hence two such caps are needed Input rms current is high in buck regulators and the worst case is when the duty cycle is 50 Duty cycle in a buck regulator can be estimated as D Vo and when this converter drives three LEDs the duty cycle will be nearly 50 Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A so two of them in parallel can tolerate more than enough for this design www national com AH90OVEIN 1 90vV IN1 LM3406 LM3406HV RECIRCULATING DIODE The input voltage of 24V 5 requires Schottky diodes with a reverse
11. is a 100 pF capacitor discharged through a 1 5 kQ resistor into each pin Note 3 Typical values represent most likely parametric norms at the conditions specified and are not guaranteed Note 4 6 of 50 C W with DAP soldered to a minimum of 2 square inches of 10z copper on the top or bottom PCB layer Note 5 Specified with junction temperature from 0 C 125 C Note 6 24V 1 1A 25 C and the load consists of three InGaN LEDs in series unless otherwise noted See the Bill of Materials table at the end of the datasheet www national com 4 Typical Performance Characteristics Efficiency Vs Number of InGaN LEDs in Series CURRENT SENSE VOLTAGE V EFFICIENCY CURRENT SENSE VOLTAGE mV Efficiency Vs Output Current Note 6 Note 6 100 _ 95 se gt Z 90 LL LL 85 80 300 600 900 1200 1500 NUMBER OF LEDS IN SERIES OUTPUT CURRENT mA Vger VS Temperature 30020363 30020364 CURRENT SENSE VOLTAGE mV 25 0 25 50 75 100 125 5 10 15 20 25 30 35 40 45 TEMPERATURE C VS LM3406HV INPUT VOLTAGE V 30020335 30020336 Current Limit vs Temperature 2 30 2 25 2 20 pa 2 10 CURRENT LIMIT A 10 20 30 40 50 60 70 80 50 25 0 25 50 75 100 125 INPUT VOLTAGE V TEMPERATURE C 30020337 30020338 5 www national com AH90OVEIN 1 90vV IN 1 LM3406 LM3406HV CURRENT LIMIT A VCC VOLTAGE V www national com
12. j Ai ton Lun 24 11 8 x 1 01 x 10 6 0 6 20 5 uH The closest standard inductor value is 22 uH The average current rating should be greater than 1 5A to prevent over heating in the inductor Inductor current ripple should be calculated for one three and five LEDs Len 24 4 1 x 5 28 x 107 22 x 10 s Leps 24 11 8 x 1 01 x 105 22 x 10 6 s Leng 24 19 7 x 1 51 x 10 22 x 10 6 The peak LED inductor current is then estimated This calcu lation uses the worst case ripple current which occurs with three LEDs lL 0 5 X Al max In order to prevent inductor saturation the inductors peak current rating must be above 1 8A A 22 uH off the shelf in ductor rated to 2 1A peak and 1 9A average with a DCR of 59 mQ will be used USING AN OUTPUT CAPACITOR This application does not require high frequency PWM dim ming allowing the use of an output capacitor to reduce the size and cost of the output inductor while still meeting the 1096 p p target for LED ripple current To select the proper output capacitor the equation from Buck Regulators with Output Ca pacitors is re arranged to yield the following Alf CIT RES X Al Alf Zc The dynamic resistance rp of one LED can be calculated by taking the tangent line to the Ve vs 1 curve in the LED datasheet Figure 8 shows an example rp calculation FORWARD VOLTAGE V 400 800 FORWAR
13. 00 us At the conclusion of this cool down time the system re starts If the current limit condition persists the cycle of cool down time and restarting will continue creating a low power hiccup mode minimizing thermal stress on the LM3406 06HV and the external circuit components OVER VOLTAGE OVER CURRENT COMPARATOR The CS pin includes an output over voltage over current comparator that will disable the power MOSFET whenever Vong exceeds 300 mV This threshold provides a hard limit for the output current Output current overshoot is limited to 300 mV Rays by this comparator during transients The OVP OCP comparator limits the maximum ripple voltage at the CS pin to 200 mV p www national com 10 OUTPUT OPEN CIRCUIT The most common failure mode for power LEDs is a broken bond wire and the result is an output open circuit When this happens the feedback path is disconnected and the output voltage will attempt to rise In buck converters the output volt age can only rise as high as the input voltage and the minimum off time requirement ensures that Vomax is slightly less than Vy Figure 4 shows a method using a zener diode Z1 and zener limiting resistor Rz to limit output voltage to the reverse breakdown voltage of Z1 plus 200 mV The zener diode reverse breakdown voltage Vz must be greater than the maximum combined V of all LEDs in the array The max imum recommended value for Rz is 1 kO The output stage SW and VOUT
14. CC loss Pg in the gate drive and linear regulator Pa lin op fsw X Qe x Vin Po 600 x 10 4 5 x 105 x 9 x 109 x 13 8 64 mW Switching loss Pg in the internal MOSFET Ps 0 5 x 13 8 1 54 x 40 x 10 9 x 4 5 x 105 190 mW AC rms current loss in the input capacitor Poin lium X ESR 0 752 0 003 2 mW negligible DCR loss in the inductor P l2 x DCR 1 542 x 0 05 120 mW Recirculating diode loss Pp 1 0 3 x 1 54 x 0 4 480 mW Current Sense Resistor Loss Pes 293 mW Electrical efficiency Sum of all loss terms 6 3 6 3 1 6 80 Temperature Rise in the LM3406 IC is calculated as m3406 Pe Pg Ps x 8j4 0 53 0 06 0 19 x 50 39 C Thermal Considerations During Input Transients The error amplifier of the LM3406 ensures that average LED current is controlled even at the transient load dump voltage of 40V leaving thermal considerations as a primary design consideration during high voltage transients A review of the operating conditions at an input of 40V is still useful to make sure that the LM3406 die temperature is not exceeded Switching frequency drops to 325 kHz the on time drops to 350 ns and the duty cycle drops to 0 12 Repeating the cal culations for conduction gate charging and switching loss leads to a total internal loss of 731 mW and hence a die tem perature rise of 37 C The LM3406 should operate properly even if
15. D CURRENT 1200 1600 2000 30020324 FIGURE 8 Calculating rp from the vs lp Curve Extending the tangent line to the ends of the plot yields values for AV and Ale of 0 7V and 2000 mA respectively Dynamic resistance is then fp AV Ale 0 7V 2A 0 350 The most filtering and therefore the highest output capaci tance is needed when rp is lowest which is when there is only one LED Inductor ripple current with one LED is 478 The required impedance of C is calculated 15 Zo 0 15 0 478 0 15 x 0 35 0 160 Aceramic capacitor will be used and the required capacitance is selected based on the impedance at 362 kHz 1 2 X TT x 0 16 x 8 62 x 105 2 75 UF This calculation assumes that Co will be a ceramic capacitor and therefore impedance due to the equivalent series resis tance ESR and equivalent series inductance ESL of of the device is negligible The closest 1096 tolerance capacitor val ue is 3 3 pF The capacitor used should be rated to 25V or more and have an X7R dielectric Several manufacturers pro duce ceramic capacitors with these specifications in the 1206 case size A typical value for ESR of 3 mQ can be read from the curve of impedance vs frequency in the product datasheet Rens Using the expression for Rays Reng 0 2 1 5 0 1330 Sub 1Q resistors are available in both 1 and 5 tolerance 1 0 130 device is the closest value and a 0 33W 1210 size
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17. National Semiconductor LM3406 06HV September 4 2008 1 5A Constant Current Buck Regulator for Driving High Power LEDs General Description The LM3406 06HV are monolithic switching regulators de signed to deliver constant currents to high power LEDs Ideal for automotive industrial and general lighting applications they contain a high side N channel MOSFET switch with a current limit of 2 0A typical for step down Buck regulators Controlled on time with true average current and an external current sense resistor allow the converter output voltage to adjust as needed to deliver a constant current to series and series parallel connected LED arrays of varying number and type LED dimming via pulse width modulation PWM is achieved using a dedicated logic pin or by PWM of the power input voltage The product feature set is rounded out with low power shutdown and thermal shutdown protection Typical Application VIN VINS RON LM3406 06HV 2008 National Semiconductor Corporation 300203 BOOT Features Integrated 2 04 MOSFET Vin Range 6V to 42V LM3406 Vin Range 6V to 75V LM3406HV True average output current control 1 7A Minimum Output Current Limit Over Temperature Cycle by Cycle Current Limit PWM Dimming with Dedicated Logic Input PWM Dimming with Power Input Voltage Simple Control Loop Compensation Low Power Shutdown Supports All Ceramic Output Capacitors and Capacitor less Outputs m The
18. ated by the following espres sion and rounding down Nmax Vo max Ve Ve forward voltage of each LED At low switching frequency Vo yax is higher allowing the LM3406 06HV to regulate output voltages that are nearly equal to input voltage and this can allow the system to drive more LEDs in series Low switching frequencies are not al ways desireable however because they require larger more expensive components CALCULATING OUTPUT VOLTAGE Even though output current is the controlled parameter in LED drivers output voltage must still be calculated in order to de sign the complete circuit Referring to the illustration in Figure 1 output voltage is calculated as Vong Sense voltage of 200 mV n number of LEDs in series MINIMUM ON TIME The minimum on time for the LM3406 06HV is 280 ns typi cal One practical example of reaching the minimum on time is when dimming the LED light output with a power FET placed in parallel to the LEDs When the FET is on the output voltage drops to 200 mV This results in a small duty cycle and in most circuits requires an on time that would be less than 280 ns In such a case the LM3406 06HV keeps the on time at 280 ns and increases the off time as much as needed which effectively reduces the switching frequency HIGH VOLTAGE BIAS REGULATOR The LM3406 06HV contains an internal linear regulator with a 7V output connected between the VIN and the VCC pins The VCC pin should be bypas
19. ching frequency when V changes so the calculation for Ron is done at the typical expected condition where Viy 13 8V and Vg 4 1V The actual Roy calculation uses the high accuracy equation listed in the Appendix 1 Ran P fsw x 1 x 10 Ron 124 kQ The closest 1 tolerance resistor is 124 kQ The switching frequency and on time of the circuit should be checked at and Vi ux Which are 9V and 16V respectively The actual few and to values have been calculated with the high accuracy equations in the Appendix 1 fou iu nei SW 4x10 x Ron 17 V lors x10 VIN ton vmax 650 ns OUTPUT INDUCTOR Since an output capacitor will be used to filter some of the LED ripple current the inductor ripple current can be set high er than the LED ripple current A value of 40 p is typical many buck converters The minimum inductance required to ensure a ripple current of 600 or less is calculated at Vi yx Vin Vo xt Ai ON L Lun 16 4 1 x 6 5 x 107 0 6 12 9 uH The closest standard inductor value is 15 pH The average current rating should be greater than 1 5A to prevent over heating in the inductor Inductor current ripple should be calculated for V y and Vin max www national com AH90OVEIN 1 90vV IN 1 LM3406 LM3406HV The peak LED inductor current is then estimated This calcu lation uses the worst case ripple current which occurs at Viy
20. d not be confused with the optical efficacy of the circuit which depends upon the LEDs themselves One calculation will be detailed for three LEDs in series where Vo 11 8V and these calculations can be repeated for other numbers of LEDs Total output power Po is calculated as lf X Vo 1 54 x 11 8 18 2W Conduction loss in the internal MOSFET www national com 16 Pc 12 Roson X D 1 542 x 0 75 x 0 5 890 mW Gate charging and VCC loss Pg in the gate drive and linear regulator Pa lin op fow X Qa x Vin 600 x 10 6 5 105 x 9 10 9 x 24 122 mW Switching loss Ps in the internal MOSFET 0 5 x 24 x 1 54 x 40 x 109 x 5 x 105 370 mW AC rms current loss Pc in the input capacitor Poin X ESR 0 752 0 003 2 mW negligible DCR loss P in the inductor P l2 x DCR 1 542 x 0 06 142 mW Recirculating diode loss Pp 1 0 5 x 1 54 x 0 4 300 mW Current Sense Resistor Loss Paus 293 mW Electrical efficiency Sum of all loss terms 18 2 18 2 2 1 89 Temperature Rise in the LM3406 IC is calculated as Po Pa Pa X 84 0 89 0 122 0 37 x 50 69 C Design Example 2 The second example circuit uses the LM3406 to drive a single white LED at 1 5 10 with a ripple current of 20 in a typical 12V automotive electrical system The two wire dim ming function will be employed in order t
21. e placed in parallel with the LED array Note that ceramic capacitors have so little ESR that it can be ignored The entire inductor ripple current still flows through Rays to provide the required 25 mV of ripple voltage for proper operation of the CS comparator AiL 30020314 FIGURE 6 LED and C Ripple Current To calculate the respective ripple currents the LED array is represented as a dynamic resistance rp LED dynamic resis tance is not always specified on the manufacturers datasheet but it can be calculated as the inverse slope of the LED s Ve vs curve Note that dividing Ve by lp will give an incorrect value that is 5x to 10x too high Total dynamic re sistance for a string of n LEDs connected in series can be calculated as the rp of one device multiplied by n Inductor ripple current is still calculated with the expression from Buck Regulators without Output Capacitors The following equa tions can then be used to estimate Ai when using a parallel Capacitor AlL Air 1 FD Zc 1 Zc 7 ESR 2n X fsw X Co The calculation for Z assumes that the shape of the inductor ripple current is approximately sinusoidal Small values of that do not significantly reduce Ai can also be used to control EMI generated by the switching action of the LM3406 06HV EMI reduction becomes more important as the length of the connections between the LED and the rest of the circuit increase INPUT CAPACITORS Input capacit
22. f the DC output voltage LED manufacturers generally recommend values for Ai ranging from 5 to 20 of Higher LED ripple current allows the use of smaller inductors smaller output capacitors or no out put capacitors at all Lower ripple current requires more output inductance higher switching frequency or additional output capacitance and may be necessary for applications that are not intended for human eyes such as machine vision or in dustrial inspection BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS The buck converter is unique among non isolated topologies because of the direct connection of the inductor to the load during the entire switching cycle By definition an inductor will control the rate of change of current that flows through it and this control over current ripple forms the basis for component selection in both voltage regulators and current regulators A current regulator such as the LED driver for which the LM3406 06HV was designed focuses on the control of the current through the load not the voltage across it A constant current regulator is free of load current transients and has no need of output capacitance to supply the load and maintain output voltage Referring to the Typical Application circuit on the front page of this datasheet the inductor and LED can form a single series chain sharing the same current When no output capacitor is used the same equations that govern inductor ripple current Ai also
23. frequency foy are related by the following expression D Vo Vp Vin Vp Vp Schottky diode typically 0 5V Vow le X Roson The LM3406 06HV regulators should be operated in contin uous conduction mode CCM where inductor current stays positive throughout the switching cycle During steady state CCM operation the converter maintains a constant switching frequency that can be estimated using the following equation for the most accurate version see the Appendix 1 fus SW 4x10 Ron SETTING LED CURRENT LED current is set by the resistor Rays which can be deter mined using the following simple expression due to the output averaging MAXIMUM NUMBER OF SERIES LEDS LED driver designers often want to determine the highest number of LEDs that can be driven by their circuits The limit on the maximum number of series LEDs is set by the highest output voltage Vo 4x that the LED driver can provide A buck regulator cannot provide an output voltage that is higher than the minimum input voltage and in pratice the maximum www national com output voltage of the LM3406 06HV is limited by the minimum off time as well Vo max determines how many LEDs can be driven in series Referring to the illustration in Figure 1 output voltage is calculated as Vo max Vinewin X 1 7 fsw X togg iN lorr ui 230 ns Once Vo max has been calculated the maximum number of series LEDs can be calcul
24. is Continuous Current Source 30020328 FIGURE 12 Parallel FET Dimming Current Loops TABLE 1 BOM for Design Example 1 DT Part Number Type Sie Parameters Oy Vendo Cin2 TABLE 2 BOM for Design Example 2 ip PatNumbe Type Size Parameters Vendor Cin1 C3225X7R1H335M Capacitor 1210 3 3 UF 50V 2 TDK Cin2 1 TK C3216X7R1H105M 1206 0 15 UF 50V 1 ERJ14RQFR13V 1210 0 130 196 CRCW08051243F 0805 124 KQ 1 Vishay 1 Panasonic CRCWO8051002F 0805 10k21 1 TABLE 3 Bill of Materials for Efficiency Curves Vishay Driver a Vishay Di Central Sem Cim Ci2 Capacitor 1812 6esp sov 2 mk Co TOK GF Cc V ob Vishay DIM1 DIM2 CS LED Vo LED 21 www national com AH90VvVEW V 90VEINT LM3406 LM3406HV Appendix The following expressions provide the best accuracy for users who wish to create computer based simulations or circuit cal culators 92 x 10 x Vo 0 65 x R pc EA AMO EI Vin 1 5 www national com 22 fsw R D fsw x 1 75 x 10 x Vin 1 5 ON 9 92 x 10 x few x Vo 0 65 Ep DAMES ME 9 92 10 x Vo 0 65 x Ron 1 75 x 10 7 x Vin 1 5 Certificate of Non Qualified Engineering Samples As a consideration for the right to sample preliminary pre production devices prior to full qualification and production release Engineering Samples by National Semiconductor Corporation incl
25. lsating currents Preference in routing should be given to the pulsat ing current paths as these are the portions of the circuit most likely to emit EMI The ground plane of a PCB is a conductor and return path and it is susceptible to noise injection just as any other circuit path The continuous current paths on the ground net can be routed on the system ground plane with less risk of injecting noise into other circuits The path be tween the input source and the input capacitor and the path between the recirculating diode and the LEDs current sense resistor are examples of continuous current paths In contrast the path between the recirculating diode and the input capac itor carries a large pulsating current This path should be routed with a short thick shape preferably on the component side of the PCB Do not place any vias near the anode of Schottky diode Instead multiple vias in parallel should be used right at the pad of the input capacitor to connect the component side shapes to the ground plane A second pul sating current loop that is often ignored is the gate drive loop formed by the SW and BOOT pins and capacitor Cg min imize this loop and the EMI it generates keep C close to the SW and BOOT pins CURRENT SENSING The CS pin is a high impedance input and the loop created by Rays Rz if used the CS pin and ground should be made www national com 20 as small as possible to maximize noise rejection Reyg sho
26. o take advantage of the legacy theater dimming method which dims and bright ens the interior lights of automobiles by chopping the input voltage with a 200Hz PWM signal As with the previous ex ample the typical Ve of a white LED is 3 9V and with the current sense voltage of 0 2V the total output voltage will be 4 1V The LED driver must operate to specifications over an input range of 9V to 16V as well as operating without suffering damage at 28V for two minutes the double battery jump start test and for 300 ms at 40V the load dump test The LED driver must also be able to operate without suffering damage at inputs as low as 6V to satisfy the cold crank tests A complete bill of materials can be found in Table 2at the end of this datasheet Vin 6V cold crank Vin 9V to 16V nominal Vin 28V 2 minutes Vin 40V 300 ms p CIN Co LM3406 Rsns 30020325 FIGURE 9 Schematic for Design Example 2 Ron and ton A switching frequency of 450 kHz helps balance the require ments of inductor size and overall power efficiency but more importantly keeps the switching frequency below 530 kHz where the AM radio band begins This design will concentrate on meeting the switching frequency and LED current require ments over the nominal input range of 9V to 16V and will then check to ensure that the transient conditions do not cause the LM3406 to overheat The LM3406 will allow a small shift in swit
27. ollowing formula can be used liN rms IF x D 1 D Ceramic capacitors are the best choice for the input to the LM3406 06HV due to their high ripple current rating low ESR low cost and small size compared to other types When se lecting a ceramic capacitor special attention must be paid to the operating conditions of the application Ceramic capaci tors can lose one half or more of their capacitance at their rated DC voltage bias and also lose capacitance with ex tremes in temperature A DC voltage rating equal to twice the expected maximum input voltage is recommended In addi 13 tion the minimum quality dielectric which is suitable for switching power supply inputs is X5R while X7R or better is preferred RECIRCULATING DIODE The LM3406 06HV is a non synchronous buck regulator that requires a recirculating diode D1 see the Typical Application circuit to carrying the inductor current during the MOSFET off time The most efficient choice for D1 is a Schottky diode due to low forward drop and near zero reverse recovery time D1 must be rated to handle the maximum input voltage plus any switching node ringing when the MOSFET is on In prac tice all switching converters have some ringing at the switch ing node due to the diode parasitic capacitance and the lead inductance D1 must also be rated to handle the average cur rent p calculated as I 1 D x I This calculation should be done at the maximum expected inpu
28. ors at the VIN pin of the LM3406 06HV are se lected using requirements for minimum capacitance and rms ripple current The input capacitors supply pulses of current approximately equal to I while the power MOSFET is on and are charged up by the input voltage while the power MOSFET is off All switching regulators have a negative input impedance due to the decrease in input current as input volt age increases This inverse proportionality of input current to input voltage can cause oscillations sometimes called power supply interaction if the magnitude of the negative input impedance is greater the the input filter impedance Minimum capacitance can be selected by comparing the input impedance to the converter s negative resistance however this requires accurate calculation of the input voltage source inductance and resistance quantities which can be difficult to determine An alternative method to select the minimum input capacitance Cin is to select the maximum input voltage ripple which can be tolerated This value Avjyyaxy S equal to the change in voltage across C during the converter on time when C supplies the load current Ci can be selected with the following P IE x ton CIN MIN AS m A good starting point for selection of Cy is to use an input voltage ripple of 5 to 10 of Vi A minimum input capaci tance of 2x the Cmn value is recommended for all LM3406 06HV circuits To determine the rms current rating the f
29. pins of the LM3406 06HV is capable of withstanding Vomax indefinitely as long as the output capacitor is rated to handle the full input voltage When an LED fails open circuit and there is no output capacitor present the surge in output voltage due to the collapsing mag netic field in the output inductor can exceed V and can damage the LM3406 06HV IC As an alternative to the zener clamp method described previously a diode can be connect ed from the output to the input of the regulator circuit that will clamp the inductive surge to one Vp above Vy D VIN VINS BOOT LM3406 6HV VOUT CS 30020311 FIGURE 4 Two Methods of Output Open Circuit Protection LOW POWER SHUTDOWN The LM3406 06HV can be placed into a low power state li sp 240 pA by grounding the RON pin with a signal level MOSFET as shown in Figure 5 Low power MOSFETs like the 2N7000 2N3904 or equivalent are recommended de vices for putting the LM3406 06HV into low power shutdown Logic gates can also be used to shut down the LM3406 06HV VIN VINS RON LM3406 06HV 2N70000r equivalent BOOT as long as the logic low voltage is below the over temperature minimum threshold of 0 3V Noise filter circuitry on the RON pin can cause a few pulses with longer on times than normal after RON is grounded or released In these cases the OVP OCP comparator will ensure that the peak inductor or LED current does not exceed 300 mV Rays VOUT CS
30. production release of such Engineer ing Samples www national com AH90OVEIN 1 90vV IN1 LM3406 LM3406HV Physical Dimensions inches millimeters unless otherwise noted 3 1 7 72 Eu 14 N un 12x 0 65 14 7 RECOMMENDED LAND TOP amp BOTTOM 6218919 RO 09 MIN ALL LEAD TIPS p 7 GAGE PLANE EXPOSED PAD 0 8 bc AT BOTTOM SEATING PLANE SEE DETAIL A 0 9 0 6 0 1 N 3 14X 0 19 0 30 0 1 0 05 IO d 0 1 CIBS ALL LEAD TIPS o 165 C BO 4 Jol 12x DIMENSIONS ARE IN MILLIMETERS 14X 0 09 0 20 MXA14A Rev A 14 Lead Exposed Pad Plastic TSSOP Package NS Package Number MXA14A www national com 24 Notes 25 www national com AH90OVEIN 1 90vV IN 1 LM3406 06HV 1 5A Constant Current Buck Regulator for Driving High Power LEDs Notes For more National Semiconductor product information and proven design tools visit the following Web sites at C NENNEN EU NENNEN NEN THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION NATIONAL PRODUCTS NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE NO LICENSE WHET
31. rent ensures that the LM3406 06HV is on when DIM pin is open circuited eliminating the need for a pull up resistor Dimming frequency 5 and duty cycle Dp are limited by the LED current rise time and fall time and the delay from activation of the DIM pin to the response of the internal power MOSFET In general should be at least one order of RON VINS DIM magnitude lower than the steady state switching frequency in order to prevent aliasing INPUT VOLTAGE COMPARATOR FOR PWM DIMMING Adding an external input diode and using the internal VINS comparator allows the LM3406 06HV to sense and respond to dimming that is done by PWM of the input voltage This method is also referred to as Two Wire Dimming and a typ ical application circuit is shown in Figure 2 If the VINS pin voltage falls 70 below the VIN pin voltage the LM3406 06HV disables the internal power FET and shuts off the current to the LED array The support circuitry driver bandgap VCC remains active in order to minimize the time needed to the turn the LED back on when the VINS pin volt age rises and exceeds 70 of VIN This minimizes the re sponse time needed to turn the LED array back on LM3406 06HV F 30020304 FIGURE 2 Typical Application using Two Wire Dimming PARALLEL MOSFET FOR HIGH SPEED PWM DIMMING For applications that require dimming at high frequency or with wide dimming duty cycle range neither the VINS com parator or the DIM pin
32. rmal Shutdown Protection m e SSOP 14 Package Applications m LED Driver Constant Current Source Automotive Lighting General Illumination Industrial Lighting VOUT CS 30020301 www national com sq31 4 104 Yang JUaIIND jue1suo WS AH9O 9OFEINT LM3406 LM3406HV Connection Diagram LM3406 06HV 1 14 2 13 3 12 4 11 5 10 6 9 7 8 30020302 14 Lead Exposed Pad Plastic TSSOP Package NS Package Number MXA14A Ordering Information Order Number Package Type NSC Package Drawing Supplied As LM3406MH 95 units in anti static rails LM3406MHX 2500 units on tape and reel eTSSOP 14 MXA14A EU DAE LM3406HVMH 95 units in anti static rails LM3406HVMHX 2500 units on tape and reel Pin Descriptions Pine Name Description Application Information Switch pin Connect these pins to the output inductor and Schottky diode BOOT MOSFET drive bootstrap pin Connect a 22 nF ceramic capacitor from this pin to the SW pins Current sense feedback pin Set the current through the LED array by connecting a resistor from this pin to ground Connect this pin to system ground Connect a logic level PWM signal to this pin to enable disable the power MOSFET and reduce the average light output of the LED array Logic high output on logic low output off www national com 2 Absolute Maximum Ratings nn posu to 0 3V to 7 LM3406 LM3406HV Note 1 RON to GND 0 3V to 7V If Mili
33. sed to the GND pin with a O 1 ceramic capacitor connected as close as possible to the pins of the IC VCC tracks VIN until VIN reaches 8 8V typical and then regulates at 7V as VIN increases The LM3406 06HV comes out of UVLO and begins operating when VCC crosses 5 3V This is shown graphically in the Typical Performance curves INTERNAL MOSFET AND DRIVER The LM3406 06HV features an internal power MOSFET as well as a floating driver connected from the SW pin to the BOOT pin Both rise time and fall time are 20 ns each typical and the approximate gate charge is 9 nC The high side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high voltage diode and an external 22 nF capacitor Cg Voc charges Cg through the internal diode while the power MOSFET is off When the MOSFET turns on the internal diode reverse biases This creates a floating supply equal to the Vcc voltage minus the diode drop to drive the MOSFET when its source voltage is equal to Viy FAST LOGIC PIN FOR PWM DIMMING The DIM pin is a TTL compatible input for PWM dimming of the LED A logic low below 0 8V at DIM will disable the in ternal MOSFET and shut off the current flow to the LED array While the DIM pin is in a logic low state the support circuitry driver bandgap VCC remains active in order to minimize the time needed to turn the LED array back on when the DIM pin sees a logic high above 2 2V A 75 pA typical pull up cur
34. stem used to control the current through an array of LEDs A dif ferential voltage signal Vans is created as the LED current flows through the current setting resistor Rays Vans is fed back by the CS pin where it is integrated and compared against an error amplifier generated reference The error am plifier is a transconductance G amplifier which adjusts the voltage on COMP to maintain a 200 mV average at the CS pin The on comparator turns on the power MOSFET when Vong falls below the reference created by the G amp The power MOSFET conducts for a controlled on time to set by an external resistor Roy the input voltage Vi and the output voltage Vo On time can be estimated by the following sim plified equation for the most accurate version of this expres sion see the Appendix Vo ton 1 x 10 x Ron x VIN At the conclusion of to the power MOSFET turns off and must remain off for a minimum of 230 ns Once this tore win is complete the CS comparator compares the integrated Vong and reference again waiting to begin the next cycle www national com AH90OVEIN 1 90vV IN 1 LM3406 LM3406HV LM3406 06HV One shot 30020306 FIGURE 1 Comparator and One Shot SWITCHING FREQUENCY The LM3406 06HV does not contain a clock however the on time is modulated in proportion to both input voltage and output voltage in order to maintain a relatively constant fre quency On time to duty cycle D and switching
35. t rating and case size The lower the duty cycle the more ther mal stress is placed on the recirculating diode When driving one LED the duty cycle can be estimated as 4 1 13 8 0 3 The estimated average diode current is then Ip 1 0 3 x 1 54 1 1A A 2A rated diode will be used To determine the proper case size the dissipation and temperature rise in D1 can be cal culated as shown in the Design Considerations section Vp for a case size such as SMB in a 60V 2A Schottky diode at 1 5A is approximately 0 4V and the 84 is 75 C W Power dis sipation and temperature rise can be calculated as Pp 1 1 x 0 4 440 mW Cc AND C The bootstrap capacitor C should always be a 22 nF ceramic capacitors with X7R dielectric A 25V rating is appropriate for all application circuits The COMP pin capacitor Cc and the linear regulator filter capacitor Ce should always be 100 nF ceramic capacitors also with X7R dielectric and a 25V rat ings EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed One calculation will be detailed for the nominal input voltage of 13 8V and these calculations can be repeat ed for other numbers of LEDs Total output power Po is calculated as Po lp x Vg 1 54 x 4 1 6 3W Conduction loss Po in the internal MOSFET Po 12 x Roson X D 1 542 x 0 75 x 0 3 530 mW Gate charging and V
36. t voltage The overall converter efficiency becomes more dependent on the selection of D1 at low duty cycles where the recirculating diode carries the load current for an increas ing percentage of the time This power dissipation can be calculating by checking the typical diode forward voltage Vp from the I V curve on the product datasheet and then multiplying it by I5 Diode datasheets will also provide a typical junction to ambient thermal resistance Oja which can be used to estimate the operating die temperature of the device Multiplying the power dissipation Pp Ip x Vp gives the temperature rise The diode case size can then be se lected to maintain the Schottky diode temperature below the operational maximum Design Example 1 The first example circuit uses the LM3406 to create a flexible LED driver capable of driving anywhere from one to five white series connected LEDs at a current of 1 5A 5 from a reg ulated DC voltage input of 24V 10 In addition to the 5 tolerance specified for the average output current the LED ripple current must be controlled to 10 of the DC value or 150 mAp_p The typical forward voltage of each individual LED at 1 5A is 3 9V hence the output voltage ranges from 4 1V to 19 7V adding in the 0 2V drop for current sensing A complete bill of materials can be found in Table 1 at the end of this datasheet www national com AH90OVEIN 1 90v IN 1 LM3406 LM3406HV Vin 24V 10
37. tary Aerospace specified devices are required Junction Temperature 150 C please contact the National Semiconductor Sales Office Storage Temp Range 65 to 125 C Distributors for availability and specifications ESD Rating Note 2 2kV VIN to GND 0 3V to 45V Soldering Information 76V LM3406HV Lead Temperature Soldering VINS to GND 0 3V to 45V 10sec 260 C 76V LM3406HV Infrared Convection Reflow 15sec 235 C VOUT to GND 0 3V to 45V u Operating Ratings BOOT to GND 0 3V to 59V 90V LM3406HV Note 1 SW to GND 1 5V to 45V Vis 6V to 42V 76V LM3406HV 75V LM3406HV BOOT to VCC 0 3V to 45V Junction Temperature Range 40 C to 125 C hdc iu nd Thermal Resistance 2 JA BOOT to SW 0 3V to 14V eTSSOP 14 Package VCC to GND 0 3V to 14V Note 4 50 C W DIM to GND 0 3V to 7V Electrical Characteristics Vn 24V unless otherwise indicated Typicals and limits appearing in plain type apply for T T 25 C Note 3 Limits appearing in boldface type apply over full Operating Temperature Range Datasheet min max specification limits are guaranteed by design test or statistical analysis LM3406 LM3406HV Symbol Parameter Conditions Win Typ Max Unit REGULATION COMPARATOR AND ERROR AMPLIFIER Veer CS Regulation Threshold CS Decreasing SW turns on 1875 200 210 191 0 210 0 lcomp COMP Pin Current Pin Current CS 0V OV UA M rn 8 i Transconductance SHUTDOWN ON AND OFF TIMER
38. the ambient temperature is as high a 85 C Layout Considerations The performance of any switching converter depends as much upon the layout of the PCB as the component selection The following guidelines will help the user design a circuit with maximum rejection of outside EMI and minimum generation of unwanted EMI COMPACT LAYOUT Parasitic inductance can be reduced by keeping the power path components close together and keeping the area of the loops that high currents travel small Short thick traces or copper pours shapes are best In particular the switch node where L1 D1 and the SW pin connect should be just large enough to connect all three components without excessive heating from the current it carries The LM3406 06HV oper ates in two distinct cycles whose high current paths are shown in Figure 11 www national com AH90OVEIN 1 90v IN 1 LM3406 LM3406HV EN 30020326 FIGURE 11 Buck Converter Current Loops The dark grey inner loop represents the high current path during the MOSFET on time The light grey outer loop rep resents the high current path during the off time GROUND PLANE AND SHAPE ROUTING The diagram of Figure 11 is also useful for analyzing the flow of continuous current vs the flow of pulsating currents The circuit paths with current flow during both the on time and off time are considered to be continuous current while those that carry current during the on time or off time only are pu
39. tringent conducted electromag netic interference tests are required Will be limited to 300 mV p or less The minimum required capacitance is calculated for the largest to 1090 ns which occurs at the minimum input voltage Using the equations from the Input Capacitors section As with the output capacitor this required value is low enough to use a ceramic capacitor and again the effective capaci tance will be lower than the rated value with 16V across Cy Reviewing plots of C vs DC Bias for several capacitors re veals that a 3 3 pF 1210 size capacitor in X7R rated to 50V loses about 22 of its rated capacitance at 16V hence two such caps are needed Input rms current is high in buck regulators and the worst case is when the duty cycle is 50 Duty cycle in a buck regulator can be estimated as D Vo Vi and when Vy drops to 9V the duty cycle will be nearly 50 Ripple current ratings for 1210 size ceramic capacitors are typically higher than 2A so two ofthem in parallel can tolerate more than enough for this design RECIRCULATING DIODE To survive an input voltage transient of 40V the Schottky diode must be rated to a higher voltage The next highest standard voltage rating is 60V Selecting a 60V rated diode provides a large safety margin for the ringing of the switch node and also makes cross referencing of diodes from differ ent vendors easier The next parameters to be determined are the forward curren
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41. uld therefore be placed as close as possible to the CS and GND pins of the IC REMOTE LED ARRAYS In some applications the LED or LED array can be far away several inches or more from the LM3406 06HV or on a sep arate PCB connected by a wiring harness When an output capacitor is used and the LED array is large or separated from the rest of the converter the output capacitor should be placed close to the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor The current sense resistor should remain on the same PCB close to the LM3406 06HV Remote LED arrays and high speed dimming with a parallel FET must be treated with special care The parallel dimming FET should be placed on the same board and or heatsink as the LEDs to minimize the loop area between them as the switching of output current by the parallel FET produces a pulsating current just like the switching action of the LM3406 s internal power FET and the Schottky diode Figure 12 shows the path that the inductor current takes through the LED or through the dimming FET To minimize the EMI from parallel FET dimming the parasitic inductance of the loop formed by the LED and the dimming FET where only the dark grey ar rows appear should be reduced as much as possible Para sitic inductance of a loop is mostly controlled by the loop area hence making this loop as physically small short as possible will reduce the inductance Buck Inductor
42. voltage rating greater than 30V The next highest standard voltage rating is 40V Selecting a 40V rated diode provides a large safety margin for the ringing of the switch node and also makes cross referencing of diodes from differ ent vendors easier The next parameters to be determined are the forward current rating and case size The lower the duty cycle the more ther mal stress is placed on the recirculating diode When driving one LED the duty cycle can be estimated as D 4 1 24 0 17 The estimated average diode current is then Ip 1 0 17 x 1 54 1 28A A 2A rated diode will be used To determine the proper case size the dissipation and temperature rise in D1 can be cal culated as shown in the Design Considerations section Vp for a case size such as SMB in a 40V 2A Schottky diode at 1 5A is approximately 0 4V and the 4 is 75 C W Power dis sipation and temperature rise can be calculated as Pp 1 28 x 0 4 512 mW Cg Cc AND The bootstrap capacitor C should always be a 22 nF ceramic capacitors with X7R dielectric A 25V rating is appropriate for all application circuits The COMP pin capacitor Cc and the linear regulator filter capacitor Ce should always be 100 nF ceramic capacitors also with X7R dielectric and a 25V rat ings EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed Electrical efficiency n shoul
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NATIONAL SEMICONDUCTOR LM3406/06HV Manual